Question 1

Ripple factor vs. form factor in rectifiers In rectifier circuits, the ripple factor RF and the form factor FF are related by which expression?

Accepted Answer

Introduction / Context: Ripple factor (RF) measures the AC ripple content relative to the DC component at a rectifier output, while form factor (FF) is the ratio of RMS value to average value of the output waveform. Their relationship is frequently tested in exams. Given Data / Assumptions: V_rms denotes RMS value of output. V_dc denotes average (DC) value of output. AC component RMS is V_ac_rms where V_ac_rms^2 = V_rms^2 − V_dc^2. Concept / Approach: By definition, RF = V_ac_rms / V_dc and FF = V_rms / V_dc. Substituting V_ac_rms in terms of V_rms and V_dc yields a direct algebraic relation between RF and FF. Step-by-Step Solution: FF = V_rms / V_dc → V_rms = FF * V_dc.V_ac_rms^2 = V_rms^2 − V_dc^2 = (FF^2 − 1) * V_dc^2.RF = V_ac_rms / V_dc = sqrt(FF^2 − 1). Verification / Alternative check: For pure DC (no ripple), FF = 1 and RF = 0, which satisfies RF = sqrt(1 − 1) = 0. Why Other Options Are Wrong: Other options contradict the squared relationship derived from RMS-DC decomposition. Common Pitfalls: Confusing average and RMS definitions or forgetting that the AC component is orthogonal to the DC component in power calculation. Final Answer: RF = (FF^2 − 1)^0.5

Question 2

Chopper (DC–DC converter) efficiency Is a DC chopper generally considered a high-efficiency power conversion circuit compared to linear regulation?

Accepted Answer

Introduction / Context: Choppers (switch-mode DC–DC converters) use high-frequency switching to adjust average voltage and current with low dissipation, unlike linear regulators which burn off the voltage difference as heat. Given Data / Assumptions: Idealized high-efficiency switches (MOSFETs/IGBTs) with low conduction and switching losses. Properly designed magnetics and filters. Continuous conduction typical but not mandatory to assert general efficiency. Concept / Approach: Switching devices operate either fully on (low voltage drop) or fully off (very low current), minimizing instantaneous power dissipation. Energy transfer via inductors/capacitors allows power processing with small losses, resulting in typical efficiencies often above 90% in practical designs. Step-by-Step Solution: Compare to linear regulation: P_loss_linear ≈ (Vin − Vout) * Iout.In a chopper, average P_loss ≪ P_out when designed well; losses are mainly switching, conduction, and magnetics.Therefore, overall efficiency is high relative to linear methods. Verification / Alternative check: Industry benchmarks for buck/boost converters routinely show 88–97% efficiency ranges under optimal conditions. Why Other Options Are Wrong: Limiting high efficiency to only low current or only high input voltage is incorrect; efficiency depends on design, not solely on these variables. Common Pitfalls: Ignoring switching and gate-drive losses at extreme frequencies or neglecting inductor core losses which can reduce efficiency if poorly selected. Final Answer: True

Question 3

Auxiliary commutated chopper (Class-C/McMurray) peak capacitor current A voltage-fed auxiliary commutated chopper uses a commutation inductor L and capacitor C with input voltage V. What is the peak capacitor current during the resonant commutation …

Accepted Answer

Introduction / Context: In auxiliary (forced) commutation of thyristor choppers, an LC network is charged and then resonated to apply a reverse current/voltage for turn-off. Estimating the peak commutating current is essential for rating the capacitor, inductor, and switching devices. Given Data / Assumptions: Ideal resonant commutation: negligible resistance during the short commutation interval. Input (charging) voltage: V. Commutation components: inductor L and capacitor C. Natural resonant frequency: ω0 = 1 / √(L * C). Concept / Approach: For a series LC excited by a step of voltage, the characteristic impedance is Z0 = √(L / C). The resulting sinusoidal current has a peak value equal to the applied step divided by Z0, provided the capacitor is driven from V to resonate with L. Step-by-Step Solution: Characteristic impedance: Z0 = √(L / C).Peak resonant current: I_peak = V / Z0.Therefore, I_peak = V / √(L / C) = V * √(C / L). Verification / Alternative check: Dimension check: √(C / L) has units of 1/Ω, so V * √(C / L) has units of ampere, consistent with current. Why Other Options Are Wrong: V / √(L * C) is voltage divided by 1/ω0, producing units of volt-second per henry (not a current); V * √(L / C) gives the inverse of the desired dependence; V * C / L has incorrect dimensions. Common Pitfalls: Confusing characteristic impedance Z0 = √(L/C) with the resonant frequency ω0; mixing up peak voltage and peak current formulas; neglecting damping, which only slightly reduces the peak in real circuits. Final Answer: V * √(C / L)

Question 4

Single-phase half-wave controlled rectifier with resistive load As the firing angle α increases, how do the DC output voltage and the RMS output voltage change?

Accepted Answer

Introduction / Context: Phase control changes the conduction interval of a controlled rectifier. For a purely resistive load in a half-wave circuit, increasing the firing angle delays conduction, thereby reducing both the average (DC) and the RMS values of the output. Given Data / Assumptions: Input: v(t) = Vm sin(ωt). Resistive load (no energy storage). Conduction over θ ∈ [α, π] each cycle. Concept / Approach: Average and RMS values are computed by integrating the output over the effective conduction interval. As α increases, the interval shortens. The RMS depends on the square of the waveform, so it does not scale linearly with the average. Step-by-Step Solution: V_dc = (Vm / 2π) * (1 − cos α) for a half-wave controlled rectifier.V_rms^2 = (1 / 2π) * ∫_α^π Vm^2 sin^2 θ dθ = (Vm^2 / 4π) * (π − α + sin 2α / 2).As α increases, both V_dc and V_rms decrease, but with different mathematical dependences (linear in 1 − cos α vs. a combination of interval and sin 2α term). Verification / Alternative check: Check limiting cases: α = 0 gives maximum V_dc and V_rms; α → π gives both tending to zero. Why Other Options Are Wrong: “Increase” options contradict the physics of delayed conduction; “same proportion” is incorrect due to differing functional forms. Common Pitfalls: Confusing average with RMS or assuming a fixed proportionality between them; forgetting the squared nature of RMS calculation. Final Answer: Both decrease (not in the same proportion)

Question 5

Series connection of SCRs (high-voltage stacks) When SCRs are connected in series for higher blocking voltage, do they require both static and dynamic equalizing (stabilizing) circuits?

Accepted Answer

Introduction / Context: Series-stacked SCRs are used to withstand higher voltages than a single device can block. Unequal device characteristics necessitate equalizing networks to avoid over-stressing any one device. Given Data / Assumptions: Multiple SCRs in series across a high DC link. Device parameter tolerances (leakage, capacitances, recovery). Presence of both steady-state and transient conditions. Concept / Approach: Static equalization (typically via resistors) ensures near-uniform voltage sharing under steady DC conditions. Dynamic equalization (typically via RC snubbers) addresses non-uniform capacitive and recovery behavior during transients (turn-off, surges). Both are necessary for reliable operation. Step-by-Step Solution: Identify problem sources: leakage mismatch (static) and junction capacitance/recovery mismatch (dynamic).Apply static resistors to bleed currents and equalize voltages at standstill.Apply RC snubbers to shape transients and share dynamic voltage stress. Verification / Alternative check: Manufacturer application notes recommend both networks for high-voltage stacks to prevent avalanche of a single device during surges. Why Other Options Are Wrong: Using only one type leaves the stack vulnerable either during steady DC (no resistors) or during switching transients (no snubbers). Common Pitfalls: Oversizing only the highest-leakage device or assuming identical parts always share stress evenly; ignoring temperature dependence of leakage. Final Answer: True

Question 6

Basic series inverter: source current waveform quality In a basic series inverter (resonant L–C with R load), the source (DC link) current generally has what harmonic content?

Accepted Answer

Introduction / Context: Series inverters rely on resonance of the L–C series branch with the load to shape current. Understanding the harmonic nature of current is crucial for filter design and source sizing. Given Data / Assumptions: Series R–L–C tuned near the fundamental of operation. Thyristor commutation via natural resonance. Quasi-sinusoidal current when tuned correctly. Concept / Approach: At or near series resonance, the impedance to the fundamental is low while it is higher for off-resonant harmonics. Thus, current tends toward a sinusoid dominated by the fundamental component, leading to lower harmonic content compared with square-wave switched currents. Step-by-Step Solution: At resonance: X_L ≈ X_C → minimum series impedance for the fundamental.Harmonics see higher impedance → attenuated current harmonics.Therefore, the source current is closer to sinusoidal and has relatively low THD. Verification / Alternative check: Practical waveforms from resonant inverters show rounded, near-sinusoidal current pulses compared to hard-switched square-wave inverters. Why Other Options Are Wrong: “High” or “zero” are extremes; properly tuned series inverters do not yield zero harmonics but typically achieve comparatively low distortion. Firing angle alone does not dictate harmonic profile; resonance tuning is key. Common Pitfalls: Assuming current is square just because devices commutate; ignoring that the series L–C shapes the current waveform strongly. Final Answer: Low harmonic content

Question 7

Choppers and commutation; Morgan chopper Assertion (A): A chopper always requires forced commutation. Reason (R): A Morgan chopper uses a saturable reactor for commutation.

Accepted Answer

Introduction / Context: DC choppers using thyristors operate on DC without natural current zero crossings. Therefore, they need a method to force devices to turn off. The Morgan chopper is a classic commutation scheme employing a saturable reactor. Given Data / Assumptions: Device: thyristor-based chopper on a DC source. No natural current zero in DC, unlike AC line commutation. Morgan commutation known as a magnetic (saturable reactor) assisted method. Concept / Approach: Since the load and source are DC, the thyristor must be reverse-biased or its current forced to zero by an auxiliary network (LC or magnetic means) to achieve turn-off. This is the essence of forced commutation. The Morgan chopper is one specific method, but the general need for forced commutation exists regardless of whether a saturable reactor is used. Step-by-Step Solution: Evaluate Assertion: For SCR-based DC choppers, forced commutation is required → True.Evaluate Reason: Morgan chopper uses a saturable reactor for commutation → True.Causality check: While R is true, it does not establish the general necessity stated in A; it merely gives one implementation example → Not a correct explanation. Verification / Alternative check: Other classes of forced commutation (Class C, D, E, F) use different networks (capacitors/inductors/pulse commutation), underscoring that forced commutation is a general need, not limited to Morgan's method. Why Other Options Are Wrong: Options suggesting A is wrong contradict basic SCR behavior on DC. Claiming R explains A would imply Morgan's method alone justifies the need, which is not the case. Common Pitfalls: Confusing “one example exists” with “general explanation”; also forgetting that with transistor switches (MOSFET/IGBT), turn-off is controlled, but the question is about SCR-based choppers. Final Answer: Both A and R are correct but R is not the correct explanation of A

Question 8

Devices with four-layer (PNPN) structures in power electronics Among the listed semiconductor devices (UJT, SCR, TRIAC), which device(s) has a four-layer PNPN structure?

Accepted Answer

Introduction / Context: Recognizing internal layer structures helps in predicting triggering behavior and conduction characteristics of power devices. In particular, four-layer PNPN structures exhibit regenerative action central to thyristor operation. Given Data / Assumptions: Options include UJT (unijunction transistor), SCR (silicon controlled rectifier), and TRIAC. We are asked which have four layers (PNPN). Concept / Approach: The SCR is the canonical four-layer device, consisting of PNPN with three junctions and three terminals (anode, cathode, gate). A TRIAC can be thought of functionally as two inverse-parallel SCRs in a single package; its physical cross-section is typically five-layer, not strictly four-layer. A UJT is a single-junction, two-layer device. Step-by-Step Solution: Identify SCR → classical PNPN (four-layer) structure.Identify TRIAC → effectively two SCRs; physical structure is generally five-layer, not a simple PNPN.Identify UJT → single junction, not PNPN.Therefore, the correct answer is SCR only. Verification / Alternative check: Device symbols and triggering models reaffirm that SCRs rely on regenerative PNPN action; UJTs and TRIACs do not match a single four-layer PNPN stack. Why Other Options Are Wrong: UJT: Single junction device; not PNPN. TRIAC: Multi-layer (often five-layer) bidirectional device, not a single PNPN stack. SCR and TRIAC: Partly incorrect because TRIAC is not strictly four-layer. Common Pitfalls: Assuming “two SCRs back-to-back” implies four layers; the integrated TRIAC structure differs. Final Answer: SCR

Question 9

Single-phase half-wave converter with resistive load (α = 0) For a single-phase half-wave controlled rectifier feeding a purely resistive load with input v = Vm sin(ωt) and firing angle α = 0°, determine the average DC output voltage Vdc and the RMS…

Accepted Answer

Introduction / Context: Basic rectifier metrics like average (DC) and RMS output voltages are foundational for sizing filters, devices, and loads. Given Data / Assumptions: Half-wave controlled rectifier, resistive load. Input v = Vm sin(ωt), firing angle α = 0° (i.e., diode-like conduction during 0 to π). Conduction interval: one half-cycle out of each period. Concept / Approach: For α = 0° and R-load, output waveform is the positive half-sine. Standard results apply: Vdc equals the average of half-sine over a full period; Vrms equals RMS of half-sine over a full period. Step-by-Step Solution: Average of half-wave rectified sine: Vdc = (1/2π) ∫₀^π Vm sinθ dθ = Vm/π.RMS of half-wave rectified sine: Vrms = √[(1/2π) ∫₀^π (Vm sinθ)^2 dθ] = Vm/2.Therefore, Vdc = Vm/π and Vrms = Vm/2. Verification / Alternative check: Numerically, for Vm = 1: Vdc ≈ 0.318, Vrms = 0.5, consistent with standard rectifier tables. Why Other Options Are Wrong: 2Vm/π or Vm/√2 pairings: These correspond to full-wave average or sinusoidal RMS, not half-wave rectified values. Common Pitfalls: Confusing full-wave with half-wave results; mixing peak and RMS of the source instead of the rectified waveform. Final Answer: Vdc = Vm/π, Vrms = Vm/2

Question 10

Peak current in an LC discharge through a thyristor path A capacitor of capacitance C is initially charged to voltage V and then discharged through a series path consisting of a thyristor, a diode, and an inductor L. What is the peak current attaine…

Accepted Answer

Introduction / Context: LC commutation and resonant discharge are common in thyristor circuits. The peak current determines device stress and component ratings. Given Data / Assumptions: Capacitor C charged to V. Series inductance L, negligible resistance. Ideal devices; energy exchange between L and C is lossless. Concept / Approach: In an ideal series LC, energy initially stored in the capacitor (E = 0.5 * C * V^2) converts entirely into inductor energy at peak current (E = 0.5 * L * I_peak^2). Equating energies yields the peak current. Step-by-Step Solution: Initial energy: E_C0 = 0.5 * C * V^2.At current peak, capacitor voltage is momentarily zero; inductor energy: E_L = 0.5 * L * I_peak^2.Set energies equal: 0.5 * C * V^2 = 0.5 * L * I_peak^2.Solve: I_peak = V * sqrt(C / L). Verification / Alternative check: Angular frequency ω₀ = 1 / sqrt(LC). Current waveform i(t) = I_peak sin(ω₀ t); peak occurs at quarter cycle, consistent with energy transfer. Why Other Options Are Wrong: V * sqrt(L/C), V * (C/L), V/L: Dimensionally incorrect or not energy-consistent. V / sqrt(L*C): Has units of current but corresponds to V * ω₀; peak current must include sqrt(C/L) factor. Common Pitfalls: Ignoring losses does not change the ideal peak expression; losses only reduce the actual peak slightly. Final Answer: V * sqrt(C/L)

Question 11

Voltage reduction due to overlap in a single-phase controlled converter A single-phase controlled converter is fed from a source with series inductance L and supplies a load current I0 (assumed constant). What is the decrease in average output volta…

Accepted Answer

Introduction / Context: Finite source inductance causes overlap between outgoing and incoming devices during commutation. This overlap reduces the average output voltage, an essential correction for accurate design. Given Data / Assumptions: Single-phase controlled converter (bridge), steady load current I0. Source inductance L; overlap angle small, sinusoidal supply. Objective is average voltage drop due to overlap. Concept / Approach: During overlap, part of the supply voltage is dropped across the source inductance while current transfers, reducing the voltage presented to the DC side. For single-phase bridges the well-known reduction is ΔV = ω L I0. Step-by-Step Solution: Inductive drop during overlap ≈ L di/dt; with constant I0 transfer over electrical angle → equivalent average drop equals ω L I0.Hence, Vdc_actual = Vdc_ideal − ω L I0. Verification / Alternative check: Detailed derivations integrating instantaneous supply minus inductive drop over the conduction interval yield the same compact correction term for small overlap. Why Other Options Are Wrong: Scaled variants (2, π, 1/π): Do not match standard single-phase result. Zero: Ignores real commutation effects of source inductance. Common Pitfalls: Confusing single-phase with three-phase results where coefficients differ; also forgetting that I0 must be essentially constant for this simple form. Final Answer: ΔV = ω L I0

Question 12

Reverse recovery time definition for a power diode In a power diode, reverse recovery time trr is measured from the instant forward current crosses zero to the instant the reverse recovery current decays to what fraction of its peak value?

Accepted Answer

Introduction / Context: Reverse recovery affects switching losses, EMI, and commutation failure risk in converters. A precise definition of trr is necessary to interpret datasheets. Given Data / Assumptions: Standard diode reverse recovery definition per common practice. Focus on the current threshold used to terminate trr measurement. Concept / Approach: After forward conduction, stored charge causes a transient reverse current when the diode is reverse-biased. The reverse recovery time is defined from forward current zero crossing to the time when reverse current falls to 10% of its peak reverse value (unless otherwise specified). Step-by-Step Solution: Identify start: i_D transitions through zero into reverse.Identify peak reverse current I_RRM.Measure time until i_D decays to 0.1 * I_RRM → trr. Verification / Alternative check: Many datasheets annotate trr using a standard test circuit and specify 10% as the end criterion for the reverse tail. Why Other Options Are Wrong: 50% / 25%: Too early; would understate trr. 0%: Unrealistic; the tail asymptotically approaches zero and measurement would be ambiguous. Common Pitfalls: Confusing trr with tq (thyristor turn-off time) or using nonstandard thresholds; always confirm the datasheet’s stated criterion. Final Answer: 10% of peak reverse current

Question 13

Input voltage vs input current waveshape in a single-phase semiconverter (R load) For a single-phase semiconverter supplying a purely resistive load, under what condition are the input voltage and input current waveshapes similar?

Accepted Answer

Introduction / Context: Power-factor and harmonic performance in controlled rectifiers are tied to how closely input current follows the input voltage. With resistive loads, conduction intervals depend on firing angle. Given Data / Assumptions: Singe-phase semiconverter (half-controlled bridge). Pure R load, no supply inductance. We compare shapes of source voltage and source current. Concept / Approach: With an R load, current is proportional to instantaneous output voltage during conduction. If α = 0°, conduction occurs during the positive half-cycle similar to a diode rectifier, giving a current waveform that mirrors the voltage half-cycle. For α > 0°, conduction begins later, so current is phase-delayed and truncated, no longer matching the voltage shape. Step-by-Step Solution: Set α = 0° → output equals the positive half-cycle of input → current is a half-sine in phase.Increase α → conduction shifts and shortens; waveform differs from the input voltage.Hence, similarity occurs only at α = 0°. Verification / Alternative check: Drawing input current for α = 0° shows an in-phase half-sine. For α = 30° or 60°, the current duration and phase shift clearly differ. Why Other Options Are Wrong: Always/Never: Too absolute; α determines similarity. High α: Produces shorter, delayed conduction—not similar. Common Pitfalls: Assuming R-load guarantees shape similarity; the firing angle changes conduction timing. Final Answer: similar only if firing angle α = 0°

Question 14

Gate pulse requirements for a single-phase full-wave AC regulator (R–L load) In a single-phase full-wave AC voltage controller feeding an R–L load with firing angle α, what is an appropriate gate-pulse strategy?

Accepted Answer

Introduction / Context: With inductive loads, current can persist after the source voltage crosses zero. Adequate gate drive ensures reliable triggering throughout the intended conduction interval. Given Data / Assumptions: Single-phase full-wave AC regulator (anti-parallel SCRs). R–L load; significant inductive behavior. Firing angle α defined from each half-cycle zero crossing. Concept / Approach: For inductive loads, short gate pulses at only the initial firing instant may be insufficient due to commutation uncertainty and device turn-on dynamics. A common practice is to apply rectangular gate pulses extending from the firing instant to near the natural turn-off, i.e., with width approximately (π − α) in each half-cycle. Step-by-Step Solution: Begin each half-cycle at ωt = 0.Trigger from ωt = α to near ωt = π to ensure robust conduction.This corresponds to rectangular pulses of width (π − α) per half-cycle. Verification / Alternative check: Practical controllers often use pulse trains or wide pulses across the expected conduction interval for R–L loads to avoid misfires. Why Other Options Are Wrong: Always short duration: Risk of mis-triggering with inductive loads. Cannot be rectangular: Incorrect; rectangular gating is common and effective. Common Pitfalls: Using ultra-short pulses suitable for R loads on R–L loads; neglecting device turn-on time and source disturbances. Final Answer: gate pulses should be rectangular with width (π − α) each half-cycle

Question 15

Current waveform in a single-phase semiconverter with highly inductive load For a single-phase semiconverter feeding a highly inductive (large L) load, what is the typical shape of the load current?

Accepted Answer

Introduction / Context: Inductive loads smooth current by storing energy. In controlled rectifiers, this affects ripple, commutation, and device stress. Given Data / Assumptions: Single-phase semiconverter (half-controlled bridge). Highly inductive load (large L), steady-state. Negligible resistance effects in shaping waveform. Concept / Approach: Large inductance resists rapid changes in current (di/dt small). Consequently, the load current becomes a slowly varying (low-ripple) waveform that, over each cycle, appears nearly flat (quasi-DC) rather than sinusoidal. Step-by-Step Solution: Apply controlled voltage pulses to an L-dominant load.Inductor stores and releases energy, smoothing current.Resultant current exhibits small ripple → nearly constant. Verification / Alternative check: Ripple magnitude ΔI ≈ (V_eff / L) * Δt; with large L, ΔI becomes small even over the conduction interval. Why Other Options Are Wrong: Sinusoidal/Rectangular: Do not reflect strong smoothing action of L. Absolutely constant: Idealized; in practice some ripple remains. Common Pitfalls: Assuming current shape follows voltage; with large L, current changes slowly and is far flatter. Final Answer: nearly constant

Question 16

Capacitor timing in an auxiliary-commutated (Class C) chopper In an auxiliary commutated chopper, how do the charging and discharging times of the commutation capacitor depend on operating conditions?

Accepted Answer

Introduction / Context: Auxiliary (Class C) choppers use a commutation capacitor and inductor to turn off the main switch. The timing of charge and discharge influences maximum switching frequency and reliability. Given Data / Assumptions: Auxiliary commutation network (C and L) is used. Main switch carries load current I_load when turned off by auxiliary path. Idealized components for reasoning. Concept / Approach: To commutate the main device, the resonant current through the auxiliary path must exceed the instantaneous load current. Thus, the discharge of the capacitor (which creates the reverse current) and the required energy in the commutation network depend on I_load at the commutation instant, altering effective timing. Step-by-Step Solution: Set resonant current i_res(t) relative to I_load.Ensure i_res_peak ≥ I_load to force current reversal in main device.Higher I_load demands more charge/discharge excursion → timing depends on load current. Verification / Alternative check: Waveforms in standard texts show commutation interval stretching with higher load current due to greater reverse current requirement. Why Other Options Are Wrong: Constant/Only source/Only inductor: Oversimplify; load current at commutation materially affects capacitor current and timing. Common Pitfalls: Designing the commutation network only for nominal current; peak-load conditions can prolong or even fail commutation if underdesigned. Final Answer: depend on load current

Question 17

Device used in relaxation oscillators for triggering and timing Which device is commonly used to build a relaxation oscillator because of its negative-resistance switching characteristic?

Accepted Answer

Introduction / Context: Relaxation oscillators generate sawtooth or pulse waveforms using a capacitor that charges through a resistor and discharges abruptly through a device with a negative-resistance region. Given Data / Assumptions: We seek the device typically used for simple, low-cost relaxation oscillators. Focus on negative-resistance triggering behavior. Concept / Approach: The unijunction transistor (UJT) has a distinct peak and valley characteristic. When the capacitor voltage reaches the peak point, the UJT conducts suddenly, discharges the capacitor to the valley point, and then turns off, producing repetitive oscillations. Step-by-Step Solution: Charge capacitor through R until V_C rises to ηV_BB + V_D.At peak point, UJT turns on → rapid discharge through emitter path.After reaching valley point, UJT turns off → cycle repeats. Verification / Alternative check: Classic lamp dimmers and SCR triggering circuits use UJT relaxation oscillators to generate timing pulses. Why Other Options Are Wrong: SCR/DIAC/TRIAC: Can participate in triggering but are not the primary negative-resistance device used alone for RC relaxation oscillators. BJT: Requires additional biasing; not inherently a relaxation device. Common Pitfalls: Confusing UJT-based oscillators with DIAC-triggered TRIAC dimmers; the UJT directly creates the relaxation action. Final Answer: UJT

Question 18

Resonant commutation vs. self-commutation in thyristor circuits What is the key difference between resonant commutation and self-commutation in power-electronic converters using L–C commutating networks?

Accepted Answer

Introduction / Context: Thyristor-based converters require a means to turn the device off, called commutation. Two widely taught forced-commutation families are resonant commutation and self-commutation. Understanding which elements carry load current is central to proper design and device stress analysis. Given Data / Assumptions: Idealized L–C components unless otherwise noted. Single-thyristor leg under discussion. Standard textbook definitions of Class A (load/resonant) and Class B (self) commutation are used. Concept / Approach: In load (resonant) commutation, an L–C network is placed so that the load current itself becomes part of a resonant loop, naturally reversing the device current and providing reverse bias. In self-commutation, a separate precharged capacitor and inductor deliver a reverse current pulse to the device without the load needing to participate in that loop. Step-by-Step Solution: Identify current paths: in resonant commutation, load current flows through L–C during the commutation interval.In self-commutation, a local L–C (or C with auxiliary path) injects reverse current into the thyristor; the load current path is largely independent during the commutation instant.Thus, the defining difference is whether the commutating elements carry the load current. Verification / Alternative check: Classic Class A inverter (load-commutated) waveforms show sinusoidal current through the series L–C and load; Class B self-commutation uses a charged capacitor across the device to force current reversal without routing the load current through the commutating elements. Why Other Options Are Wrong: (a) They are not “entirely different”; both exploit L–C resonance. (c) Self-commutation typically employs L and C, not only C. (d) Incorrect because (b) is correct. (e) Resonant commutation explicitly uses reactive elements. Common Pitfalls: Confusing “load commutation” with “line commutation”. Assuming any capacitor turn-off is “self-commutation” without checking current paths. Final Answer: In resonant commutation, the commutating elements carry the load current, whereas in self-commutation the commutating elements do not carry the load current

Question 19

Single-phase cycloconverter structure For single-phase operation, how many converter groups does a cycloconverter use?

Accepted Answer

Introduction / Context: Cycloconverters synthesize a lower-frequency AC output directly from an AC input by selecting and modulating segments of the input waveform. In single-phase designs, bidirectional power flow at the output requires two controlled rectifier groups. Given Data / Assumptions: Single-phase input and single-phase output. Conventional configuration with anti-parallel converter groups. Concept / Approach: To generate both positive and negative halves of the low-frequency output, one converter group (the positive group) conducts during positive output half-cycles, while the other (the negative group) conducts during negative output half-cycles. Step-by-Step Solution: Output polarity alternates at the synthesized low frequency.Provide a controlled rectifier group dedicated to each polarity: one for +ve output, one for −ve output.Hence, exactly one positive group and one negative group are required. Verification / Alternative check: Standard block diagrams of single-phase cycloconverters show two controlled bridges connected in anti-parallel feeding the same load. Why Other Options Are Wrong: (b), (c) Add unnecessary redundancy for single-phase output. (d) Not true—group count is specific. (e) Practical implementations separate groups for gating simplicity and device stress control. Common Pitfalls: Confusing single-phase with three-phase cycloconverter structures. Assuming a single bridge can cleanly synthesize both polarities without reversed connections. Final Answer: One positive-group and one negative-group converter

Question 20

Commutation classes and “two-lead” terminology Which commutation method is commonly referred to as using two leads (two main thyristors commutating each other)?

Accepted Answer

Introduction / Context: Textbook thyristor commutation methods are labeled Class A through Class F. The term “two-lead” or “complementary commutation” typically refers to arrangements where two main SCRs aid in commutating each other via a shared commutation network. Given Data / Assumptions: Naming follows standard power electronics literature. Focus on single-quadrant choppers/inverters nomenclature. Concept / Approach: In Class C (complementary) commutation, two main thyristors are engaged so that turning one on helps reverse-bias and turn the other off. This pairing is the typical basis for the “two-lead” descriptor. Step-by-Step Solution: Identify complementary action: each device provides the path to remove charge from the other.Network elements are arranged so conduction of one device assists the forced turn-off of its complement.Therefore, the method aligned with the “two-lead” naming is Class C. Verification / Alternative check: Many references list Class C as complementary/two-thyristor commutation used in choppers and inverters. Why Other Options Are Wrong: Class A is load (resonant) commutation. Class B is self (resonant-pulse) commutation. Class D is auxiliary-pulse commutation; Class E/F are specialized. Common Pitfalls: Mixing “two-lead commutation” with “two-quadrant choppers”; they are related but not identical terms. Final Answer: Class C

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