Q: Wave velocity in a dielectric-filled cable A uniform cable has relative permittivity ε_r = 4 (μ_r ≈ 1). What is the phase velocity of electromagnetic waves in this cable?

Introduction / Context: Phase velocity in a medium directly affects signal delay, line length for a given electrical length, and dispersion analysis. Designers regularly convert between free-space parameters and substrate/cable parameters using ε_r and μ_r. Given Data / Assumptions: Relative permittivity ε_r = 4. Relative permeability μ_r ≈ 1 (nonmagnetic dielectric). Free-space speed of light c = 3 × 10^8 m/s. Concept / Approach: The phase velocity in a dielectric is v = c / √(ε_r μ_r). With μ_r ≈ 1, v = c / √ε_r. Substituting ε_r = 4 gives √ε_r = 2, hence v = c/2 = 1.5 × 10^8 m/s. This result is fundamental to transmission-line timing and length calculations in RF and high-speed digital designs. Step-by-Step Solution: 1) Formula: v = c / √(ε_r μ_r) → with μ_r = 1, v = c / √ε_r.2) Compute √ε_r = √4 = 2.3) v = 3 × 10^8 / 2 = 1.5 × 10^8 m/s. Verification / Alternative check: Velocity factor VF = v/c = 1/√ε_r = 0.5. Many cables with ε_r ≈ 4 exhibit VF ≈ 0.5, confirming the calculation. Why Other Options Are Wrong: 3 × 10^8 m/s corresponds to air (ε_r = 1). 0.75 × 10^8 and 1 × 10^8 m/s do not satisfy v = c/√4. 2.25 × 10^8 m/s equates to ε_r ≈ 1.78, not 4. Common Pitfalls: Mixing phase and group velocities or forgetting to take the square root of ε_r. Final Answer: 1.5 × 10^8 m/s.

Q: Microwave resonator Q factors – Relationship between loaded Q, external Q, and unloaded Q For a single-resonator coupled system, relate the loaded quality factor QL, the external (coupling) quality factor QE, and the unloaded (intrinsic) quality fac…

Introduction / Context: Quality factors quantify energy storage versus loss in resonators. In practical microwave filters and oscillators, a resonator is coupled to external circuitry, so the observed (loaded) Q includes both intrinsic losses and coupling losses. Knowing how these combine is critical for bandwidth, insertion loss, and sensitivity calculations. Given Data / Assumptions: Single resonator coupled to a port or network. Intrinsic loss modeled by QU; coupling loss modeled by QE. Loaded resonance includes both loss mechanisms. Concept / Approach: The rate of energy loss adds in terms of 1/Q. Therefore the standard relation is 1/QL = 1/QU + 1/QE. Solving for QL gives QL = (QE * QU) / (QE + QU). This expression directly relates coupling strength (QE) to the observed 3 dB bandwidth and informs design trade-offs between selectivity and insertion loss. Step-by-Step Solution: 1) Combine losses: 1/QL = 1/QU + 1/QE.2) Invert both sides: QL = 1 / (1/QU + 1/QE).3) Simplify: QL = (QE * QU) / (QE + QU). Verification / Alternative check: Limiting cases: (i) Weak coupling QE → ∞ ⇒ QL → QU. (ii) Strong coupling QE ≪ QU ⇒ QL ≈ QE. Both limits match intuition, confirming correctness. Why Other Options Are Wrong: Option A is an intermediate relation but is not the requested expression for QL; option B and C are incorrect arithmetic combinations; option E is not physically derived for additive loss channels. Common Pitfalls: Confusing series vs. parallel addition of Qs; mixing bandwidth definitions without relating them back to Q via Q = f0 / Δf. Final Answer: QL = (QE * QU) / (QE + QU).

Q: Microwaves for communication – Key advantage What is the principal advantage of using microwave frequencies for communication links, compared with lower-frequency radio bands?

Introduction / Context: Microwave spectrum allocations provide wide contiguous bandwidths, which is the central driver for high-capacity links such as satellite backhaul, point-to-point radios, and modern WLANs. Bandwidth is the currency of data rate, per Shannon’s capacity relation and practical modulation schemes. Given Data / Assumptions: Comparing microwaves (roughly 1–30 GHz) with lower-frequency HF/VHF bands. Considering typical regulatory allocations and available channel widths. Assuming appropriate link budget and line-of-sight where required. Concept / Approach: Wider bandwidth directly supports higher symbol rates and therefore higher throughput using QAM/OFDM. While microwaves have higher free-space loss and may suffer rain fade, the sheer bandwidth availability outweighs these disadvantages for many systems, especially with high-gain antennas and spatial reuse (narrow beams reduce interference and enable frequency reuse). Step-by-Step Solution: 1) Identify need: high data rate → requires large bandwidth.2) Microwave bands offer multi-MHz to multi-GHz channels vs. tens of kHz at lower bands.3) Apply capacity intuition: higher B supports higher R for given SNR and modulation. Verification / Alternative check: Real systems: Wi-Fi at 2.4/5/6 GHz, E-band (70/80 GHz) backhaul offering multi-Gb/s, and Ku/Ka satellite links exemplify bandwidth-driven capacity at microwaves. Why Other Options Are Wrong: Options C and D confuse power with advantage; E is false because atmospheric effects can be significant at some microwave bands; B contradicts the capacity objective. Common Pitfalls: Assuming path loss alone dictates feasibility; in practice, high-gain dishes and narrow beams make microwave links practical and spectrum-efficient. Final Answer: large available bandwidth enabling high data rates.

Q: Understanding TE10 in a rectangular waveguide Which statement best describes the physical nature of TE10 propagation in a hollow rectangular waveguide?

Introduction / Context: Waveguides do not support TEM modes unless multiple conductors are present. Rectangular waveguides are single-conductor enclosures; their propagating solutions are TE or TM, each with cutoff. The TE10 mode is the dominant mode in a standard rectangular waveguide and is a genuine TE solution to Maxwell’s equations with boundary conditions, not a superposition of TEM modes. Given Data / Assumptions: Single hollow PEC rectangular guide, air-filled. TE10 is dominant with cutoff f_c = c/(2a). No center conductor → no true TEM mode exists. Concept / Approach: TEM waves require at least two conductors to define a transverse E field with zero longitudinal components everywhere. In a single-conductor hollow guide, boundary conditions enforce solutions with nonzero longitudinal components (either Ez or Hz). TE10 has Hz ≠ 0 with Ez = 0, exhibiting a standing variation across the broad wall and propagating along z with phase constant β above cutoff. Step-by-Step Solution: 1) Recognize geometry → single conductor → no pure TEM mode.2) Solve for fields → TE and TM families only.3) TE10 arises from transverse boundary conditions; not expressible as a finite sum of TEM modes. Verification / Alternative check: Mode charts and derivations show Ez = 0 for TE modes, Ez ≠ 0 for TM modes, and neither equals the TEM condition (Ez = Hz = 0). Why Other Options Are Wrong: Options A–C attempt to reinterpret TE10 as combinations of TEM, which is physically incorrect in a single-conductor guide. Common Pitfalls: Assuming “all waves are TEM” or importing transmission-line intuition to waveguides without considering boundary conditions and cutoff behavior. Final Answer: None of the above descriptions are correct because TE10 is a true TE mode, not a combination of TEM waves.

Q: Assertion–Reason: Artificial transmission line as a pulse generator Assertion (A): An artificial (lumped) transmission line can be used as a pulse generator (pulse-forming network). Reason (R): A pulse function is used very frequently in network ana…

Introduction / Context: Artificial transmission lines (pulse-forming networks, PFNs) are cascaded L–C sections that emulate the delay and impedance of a real line. When charged and discharged through a switch into a matched load, they generate rectangular pulses with controlled width and amplitude, widely used in radar modulators and pulsed power systems. Given Data / Assumptions: An artificial line approximates Z0 and propagation delay with lumped L and C. When properly switched into Z0, it produces a flat-top pulse until the stored energy is depleted. Network analysis often employs pulse or step functions for system characterization. Concept / Approach: The validity of A: A PFN charged to V and suddenly connected to a matched load Z0 discharges to produce a pulse approximately of amplitude V/2 (for certain topologies) and duration determined by the line’s electrical length. This is a standard pulse generation method independent of any analytic preference. The truth of R: Pulse functions are indeed used frequently for analysis, but that fact does not explain why an artificial line works as a pulse generator; the explanation rests on energy storage and transmission-line matching, not on mathematical usage of pulses. Step-by-Step Solution: 1) Synthesize an L–C ladder with Z0 and delay τ.2) Charge to V using a DC source.3) Close a switch to a matched load → pulse of width ≈ 2τ and controlled amplitude. Verification / Alternative check: Time-domain analysis of ladder networks shows uniform current/voltage segments during discharge into Z0, matching pulse-forming behavior. Why Other Options Are Wrong: Option A: claims R explains A, which is incorrect causality; options C/D/E deny one or both truths contrary to standard PFN practice. Common Pitfalls: Confusing pulse analysis with pulse generation; overlooking the requirement of matching the load to Z0 for a flat pulse. Final Answer: Both A and R are correct but R is not the correct explanation of A.

Q: Power monitoring and reflectometry – Underlying principle Modern microwave power monitors and reflectometers primarily rely on which fundamental microwave component principle?

Introduction / Context: Measuring forward and reflected power on transmission lines requires separating the two traveling waves. Directional couplers provide this separation, enabling direct sampling of forward and reverse waves for power monitoring and VSWR/return-loss measurement in reflectometers. Given Data / Assumptions: Coupled-line or waveguide directional couplers are available. Detectors or power sensors can be attached to coupled ports. Goal is to measure forward and/or reflected power accurately. Concept / Approach: A directional coupler couples a known fraction of power from the main line into auxiliary ports with a preferred directionality. By attaching detectors to the forward and reverse coupled ports, instruments compute power, return loss, and VSWR. This principle underpins reflectometers and most in-line wattmeters used in microwave labs and systems. Step-by-Step Solution: 1) Insert a directional coupler with specified coupling C dB and directivity D dB.2) Connect detectors to forward and reverse coupled ports.3) Convert detected voltages to powers, then compute reflection coefficient and VSWR. Verification / Alternative check: Calibrated couplers with known coupling enable absolute power estimation; dual-directional couplers permit simultaneous forward and reverse readings for reflectometry. Why Other Options Are Wrong: Klystrons, Gunn diodes, and TWTs are sources/amplifiers, not measuring elements; a DRO is an oscillator, not a wave-separating device. Common Pitfalls: Confusing coupling with insertion loss; ignoring directivity limits which set the floor for measurement accuracy in high-return-loss systems. Final Answer: directional coupler.

Q: SWR → ∞ conditions on a transmission line For a lossless line with characteristic impedance Z0, the standing-wave ratio (SWR) becomes infinite when |Γ| = 1. Which load terminations guarantee |Γ| = 1? Select the correct combination:

Introduction / Context: SWR quantifies how severe standing waves are due to reflection from a load. Infinite SWR corresponds to complete reflection (|Γ| = 1). This occurs for specific classes of terminations and has direct implications for power handling and node voltages on lines. Given Data / Assumptions: Lossless line with characteristic impedance Z0 (real > 0). Loads considered: (1) short circuit, (2) arbitrary complex impedance, (3) open circuit, (4) purely reactive load jX. Standard definitions: Γ = (ZL − Z0) / (ZL + Z0), VSWR = (1 + |Γ|) / (1 − |Γ|). Concept / Approach: |Γ| = 1 whenever the load is lossless and not equal to Z0 in magnitude/phase such that the denominator magnitude equals the numerator magnitude. Shorts (ZL = 0) and opens (ZL → ∞) obviously yield |Γ| = 1. Purely reactive loads ZL = jX also give |Γ| = 1 because |jX − Z0| = |jX + Z0| for real Z0, leading to unit magnitude reflection (phase depends on X). A general “complex impedance” may have |Γ| < 1 if it possesses a resistive part not equal to 0 and not infinite; therefore it does not guarantee |Γ| = 1. Step-by-Step Solution: 1) Short: Γ = (0 − Z0)/(0 + Z0) = −1 → |Γ| = 1.2) Open: Γ → +1 → |Γ| = 1.3) Pure reactance: Γ = (jX − Z0)/(jX + Z0) → |Γ| = 1 (magnitudes equal). Verification / Alternative check: Compute VSWR from |Γ|; |Γ| = 1 implies denominator 0 → SWR → ∞, matching intuition of complete reflection. Why Other Options Are Wrong: Any option including “complex impedance” as a guarantee is overbroad; many complex impedances (with resistive parts) yield |Γ| < 1. Options excluding short/open/pure reactance miss guaranteed |Γ| = 1 cases. Common Pitfalls: Believing any non-matched impedance leads to |Γ| = 1; in reality, |Γ| spans (0, 1] depending on mismatch severity. Final Answer: 1, 3, and 4 (short circuit; open circuit; pure reactance).

Q: Rectangular Waveguide Cutoff Relation (Dominant TE10) A rectangular waveguide has broad-wall width a. At cutoff for the dominant TE10 mode, which relation between a and the cutoff wavelength λc is correct?

Introduction / Context: Cutoff conditions in rectangular waveguides determine the lowest frequency that can propagate for a given mode. For the dominant TE10 mode, the broad dimension a controls the cutoff wavelength and hence the usable frequency band for single-mode operation. Given Data / Assumptions: Uniform, air-filled rectangular waveguide with width a (broad wall) and height b (narrow wall). Dominant mode is TE10, with one half-sine variation across a and none across b. Perfect electric conductor approximation for walls. Concept / Approach: The cutoff wavelength for mode TEmn in a rectangular guide is λc = 2 / sqrt((m/a)^2 + (n/b)^2). For TE10, m = 1 and n = 0, giving λc = 2a. Rearranging yields a = λc / 2. This simple proportionality is a foundational design rule for waveguides and is frequently used for selecting a standard WR size given a desired band. Step-by-Step Solution: 1) Start from general formula: λc = 2 / sqrt((1/a)^2 + 0) = 2a.2) Rearrange: a = λc / 2.3) Therefore, at TE10 cutoff, the broad-wall width equals half of the cutoff wavelength. Verification / Alternative check: The well-known dominant-mode cutoff frequency is fc10 = c / (2a). Converting to wavelength (λc = c / fc10) again gives λc = 2a, consistent with the above. Why Other Options Are Wrong: a = λc or a = 2λc contradict the derived relation; a = λc / 4 is also incorrect; “independent of λc” ignores the defining cutoff equation. Common Pitfalls: Confusing λ (operating wavelength) with λc (cutoff wavelength); λ must be shorter than λc for propagation in a given mode. Final Answer: a = λc / 2

Question 1

Rectangular waveguide cutoff relationship In a rectangular waveguide, the cutoff wavelength for the dominant TE10 mode is measured as 8 cm. Determine the cutoff wavelength for the TE20 mode of the same waveguide, assuming standard air-filled operati…

Accepted Answer

Introduction / Context: The question checks understanding of how cutoff wavelengths scale with mode indices in rectangular waveguides. Knowing the geometry-to-cutoff relationship lets engineers quickly infer the behavior of higher-order modes once the dominant mode is known. Given Data / Assumptions: Air-filled rectangular waveguide (μ_r ≈ 1, ε_r ≈ 1). Cutoff wavelength for TE10 is given as 8 cm. Standard rectangular guide dimensions: broad wall a, narrow wall b; TE_mn cutoff depends on a and b. Concept / Approach: For a rectangular waveguide, the cutoff wavelength for TE_mn modes is λ_c(mn) = 2 / √((m/a)^2 + (n/b)^2). For TE10, λ_c10 = 2a. Hence, if λ_c10 is known, we can compute a and then evaluate λ_c for other modes. For TE20, λ_c20 = a (since m = 2 gives 2 / (2/a) = a). Step-by-Step Solution: 1) From λ_c10 = 2a = 8 cm → a = 4 cm.2) TE20 cutoff wavelength: λ_c20 = a = 4 cm.3) No dependence on b for TE20 because n = 0 → term in b vanishes. Verification / Alternative check: Scaling logic: doubling the index along the a-dimension halves the cutoff wavelength (TE10 to TE20 halves λ_c). 8 cm → 4 cm confirms consistency. Why Other Options Are Wrong: 8 cm assumes same mode; 6 cm and 2 cm do not match the exact TE_mn formula for m = 2; 12 cm would be larger than the dominant mode, which is impossible for higher-order modes. Common Pitfalls: Confusing TE20 with TE01 (which involves b), or thinking cutoff scales with physical length rather than modal indices. Final Answer: 4 cm.

Question 2

Coaxial microwave line – Practical single-mode operating range For a coaxial transmission line (supporting TEM as the fundamental mode), at what frequency range should normal operation be restricted to ensure single-mode propagation and avoid higher…

Accepted Answer

Introduction / Context: Coaxial lines support a TEM fundamental mode that has no cutoff, but they also support higher-order modes (like TE11) above specific cutoff frequencies. For low-distortion, predictable measurements and links, designers must keep operation in the single-mode region. Given Data / Assumptions: Standard coaxial geometry with inner and outer conductors. Desire is to avoid multimode propagation. TEM mode is desired; TE11 is the first higher-order mode with a finite cutoff. Concept / Approach: Although the TEM mode has no cutoff, once the frequency approaches the TE11 cutoff, fields can couple to TE11. Above that cutoff, the line can carry multiple modes, causing dispersion, mismatch errors, and unpredictable coupling. Therefore, the usable range is constrained to frequencies below the TE11 cutoff (often with margin), ensuring only the TEM mode propagates. Step-by-Step Solution: 1) Identify first higher-order mode in coax → TE11.2) Determine its cutoff frequency f_c,TE11 from geometry.3) Restrict operation to f < f_c,TE11 (ideally f ≤ 0.7–0.8 f_c as engineering margin). Verification / Alternative check: Vendor datasheets specify a maximum single-mode frequency for each connector/line size, always below the TE11 cutoff, validating the rule of thumb. Why Other Options Are Wrong: Operating above or at cutoff risks TE11 excitation; “much higher than cutoff” ensures multimode propagation; “much lower” (option C) is overly restrictive wording not required by the question. Common Pitfalls: Confusing “no cutoff” of TEM with “unlimited bandwidth.” In practice, higher modes and connectors limit usable bandwidth. Final Answer: below the cutoff frequency of the TE11 mode.

Question 3

Wave velocity in a dielectric-filled cable A uniform cable has relative permittivity ε_r = 4 (μ_r ≈ 1). What is the phase velocity of electromagnetic waves in this cable?

Accepted Answer

Introduction / Context: Phase velocity in a medium directly affects signal delay, line length for a given electrical length, and dispersion analysis. Designers regularly convert between free-space parameters and substrate/cable parameters using ε_r and μ_r. Given Data / Assumptions: Relative permittivity ε_r = 4. Relative permeability μ_r ≈ 1 (nonmagnetic dielectric). Free-space speed of light c = 3 × 10^8 m/s. Concept / Approach: The phase velocity in a dielectric is v = c / √(ε_r μ_r). With μ_r ≈ 1, v = c / √ε_r. Substituting ε_r = 4 gives √ε_r = 2, hence v = c/2 = 1.5 × 10^8 m/s. This result is fundamental to transmission-line timing and length calculations in RF and high-speed digital designs. Step-by-Step Solution: 1) Formula: v = c / √(ε_r μ_r) → with μ_r = 1, v = c / √ε_r.2) Compute √ε_r = √4 = 2.3) v = 3 × 10^8 / 2 = 1.5 × 10^8 m/s. Verification / Alternative check: Velocity factor VF = v/c = 1/√ε_r = 0.5. Many cables with ε_r ≈ 4 exhibit VF ≈ 0.5, confirming the calculation. Why Other Options Are Wrong: 3 × 10^8 m/s corresponds to air (ε_r = 1). 0.75 × 10^8 and 1 × 10^8 m/s do not satisfy v = c/√4. 2.25 × 10^8 m/s equates to ε_r ≈ 1.78, not 4. Common Pitfalls: Mixing phase and group velocities or forgetting to take the square root of ε_r. Final Answer: 1.5 × 10^8 m/s.

Question 4

Microwave resonator Q factors – Relationship between loaded Q, external Q, and unloaded Q For a single-resonator coupled system, relate the loaded quality factor QL, the external (coupling) quality factor QE, and the unloaded (intrinsic) quality fac…

Accepted Answer

Introduction / Context: Quality factors quantify energy storage versus loss in resonators. In practical microwave filters and oscillators, a resonator is coupled to external circuitry, so the observed (loaded) Q includes both intrinsic losses and coupling losses. Knowing how these combine is critical for bandwidth, insertion loss, and sensitivity calculations. Given Data / Assumptions: Single resonator coupled to a port or network. Intrinsic loss modeled by QU; coupling loss modeled by QE. Loaded resonance includes both loss mechanisms. Concept / Approach: The rate of energy loss adds in terms of 1/Q. Therefore the standard relation is 1/QL = 1/QU + 1/QE. Solving for QL gives QL = (QE * QU) / (QE + QU). This expression directly relates coupling strength (QE) to the observed 3 dB bandwidth and informs design trade-offs between selectivity and insertion loss. Step-by-Step Solution: 1) Combine losses: 1/QL = 1/QU + 1/QE.2) Invert both sides: QL = 1 / (1/QU + 1/QE).3) Simplify: QL = (QE * QU) / (QE + QU). Verification / Alternative check: Limiting cases: (i) Weak coupling QE → ∞ ⇒ QL → QU. (ii) Strong coupling QE ≪ QU ⇒ QL ≈ QE. Both limits match intuition, confirming correctness. Why Other Options Are Wrong: Option A is an intermediate relation but is not the requested expression for QL; option B and C are incorrect arithmetic combinations; option E is not physically derived for additive loss channels. Common Pitfalls: Confusing series vs. parallel addition of Qs; mixing bandwidth definitions without relating them back to Q via Q = f0 / Δf. Final Answer: QL = (QE * QU) / (QE + QU).

Question 5

Microwaves for communication – Key advantage What is the principal advantage of using microwave frequencies for communication links, compared with lower-frequency radio bands?

Accepted Answer

Introduction / Context: Microwave spectrum allocations provide wide contiguous bandwidths, which is the central driver for high-capacity links such as satellite backhaul, point-to-point radios, and modern WLANs. Bandwidth is the currency of data rate, per Shannon’s capacity relation and practical modulation schemes. Given Data / Assumptions: Comparing microwaves (roughly 1–30 GHz) with lower-frequency HF/VHF bands. Considering typical regulatory allocations and available channel widths. Assuming appropriate link budget and line-of-sight where required. Concept / Approach: Wider bandwidth directly supports higher symbol rates and therefore higher throughput using QAM/OFDM. While microwaves have higher free-space loss and may suffer rain fade, the sheer bandwidth availability outweighs these disadvantages for many systems, especially with high-gain antennas and spatial reuse (narrow beams reduce interference and enable frequency reuse). Step-by-Step Solution: 1) Identify need: high data rate → requires large bandwidth.2) Microwave bands offer multi-MHz to multi-GHz channels vs. tens of kHz at lower bands.3) Apply capacity intuition: higher B supports higher R for given SNR and modulation. Verification / Alternative check: Real systems: Wi-Fi at 2.4/5/6 GHz, E-band (70/80 GHz) backhaul offering multi-Gb/s, and Ku/Ka satellite links exemplify bandwidth-driven capacity at microwaves. Why Other Options Are Wrong: Options C and D confuse power with advantage; E is false because atmospheric effects can be significant at some microwave bands; B contradicts the capacity objective. Common Pitfalls: Assuming path loss alone dictates feasibility; in practice, high-gain dishes and narrow beams make microwave links practical and spectrum-efficient. Final Answer: large available bandwidth enabling high data rates.

Question 6

Understanding TE10 in a rectangular waveguide Which statement best describes the physical nature of TE10 propagation in a hollow rectangular waveguide?

Accepted Answer

Introduction / Context: Waveguides do not support TEM modes unless multiple conductors are present. Rectangular waveguides are single-conductor enclosures; their propagating solutions are TE or TM, each with cutoff. The TE10 mode is the dominant mode in a standard rectangular waveguide and is a genuine TE solution to Maxwell’s equations with boundary conditions, not a superposition of TEM modes. Given Data / Assumptions: Single hollow PEC rectangular guide, air-filled. TE10 is dominant with cutoff f_c = c/(2a). No center conductor → no true TEM mode exists. Concept / Approach: TEM waves require at least two conductors to define a transverse E field with zero longitudinal components everywhere. In a single-conductor hollow guide, boundary conditions enforce solutions with nonzero longitudinal components (either Ez or Hz). TE10 has Hz ≠ 0 with Ez = 0, exhibiting a standing variation across the broad wall and propagating along z with phase constant β above cutoff. Step-by-Step Solution: 1) Recognize geometry → single conductor → no pure TEM mode.2) Solve for fields → TE and TM families only.3) TE10 arises from transverse boundary conditions; not expressible as a finite sum of TEM modes. Verification / Alternative check: Mode charts and derivations show Ez = 0 for TE modes, Ez ≠ 0 for TM modes, and neither equals the TEM condition (Ez = Hz = 0). Why Other Options Are Wrong: Options A–C attempt to reinterpret TE10 as combinations of TEM, which is physically incorrect in a single-conductor guide. Common Pitfalls: Assuming “all waves are TEM” or importing transmission-line intuition to waveguides without considering boundary conditions and cutoff behavior. Final Answer: None of the above descriptions are correct because TE10 is a true TE mode, not a combination of TEM waves.

Question 7

Assertion–Reason: Artificial transmission line as a pulse generator Assertion (A): An artificial (lumped) transmission line can be used as a pulse generator (pulse-forming network). Reason (R): A pulse function is used very frequently in network ana…

Accepted Answer

Introduction / Context: Artificial transmission lines (pulse-forming networks, PFNs) are cascaded L–C sections that emulate the delay and impedance of a real line. When charged and discharged through a switch into a matched load, they generate rectangular pulses with controlled width and amplitude, widely used in radar modulators and pulsed power systems. Given Data / Assumptions: An artificial line approximates Z0 and propagation delay with lumped L and C. When properly switched into Z0, it produces a flat-top pulse until the stored energy is depleted. Network analysis often employs pulse or step functions for system characterization. Concept / Approach: The validity of A: A PFN charged to V and suddenly connected to a matched load Z0 discharges to produce a pulse approximately of amplitude V/2 (for certain topologies) and duration determined by the line’s electrical length. This is a standard pulse generation method independent of any analytic preference. The truth of R: Pulse functions are indeed used frequently for analysis, but that fact does not explain why an artificial line works as a pulse generator; the explanation rests on energy storage and transmission-line matching, not on mathematical usage of pulses. Step-by-Step Solution: 1) Synthesize an L–C ladder with Z0 and delay τ.2) Charge to V using a DC source.3) Close a switch to a matched load → pulse of width ≈ 2τ and controlled amplitude. Verification / Alternative check: Time-domain analysis of ladder networks shows uniform current/voltage segments during discharge into Z0, matching pulse-forming behavior. Why Other Options Are Wrong: Option A: claims R explains A, which is incorrect causality; options C/D/E deny one or both truths contrary to standard PFN practice. Common Pitfalls: Confusing pulse analysis with pulse generation; overlooking the requirement of matching the load to Z0 for a flat pulse. Final Answer: Both A and R are correct but R is not the correct explanation of A.

Question 8

Power monitoring and reflectometry – Underlying principle Modern microwave power monitors and reflectometers primarily rely on which fundamental microwave component principle?

Accepted Answer

Introduction / Context: Measuring forward and reflected power on transmission lines requires separating the two traveling waves. Directional couplers provide this separation, enabling direct sampling of forward and reverse waves for power monitoring and VSWR/return-loss measurement in reflectometers. Given Data / Assumptions: Coupled-line or waveguide directional couplers are available. Detectors or power sensors can be attached to coupled ports. Goal is to measure forward and/or reflected power accurately. Concept / Approach: A directional coupler couples a known fraction of power from the main line into auxiliary ports with a preferred directionality. By attaching detectors to the forward and reverse coupled ports, instruments compute power, return loss, and VSWR. This principle underpins reflectometers and most in-line wattmeters used in microwave labs and systems. Step-by-Step Solution: 1) Insert a directional coupler with specified coupling C dB and directivity D dB.2) Connect detectors to forward and reverse coupled ports.3) Convert detected voltages to powers, then compute reflection coefficient and VSWR. Verification / Alternative check: Calibrated couplers with known coupling enable absolute power estimation; dual-directional couplers permit simultaneous forward and reverse readings for reflectometry. Why Other Options Are Wrong: Klystrons, Gunn diodes, and TWTs are sources/amplifiers, not measuring elements; a DRO is an oscillator, not a wave-separating device. Common Pitfalls: Confusing coupling with insertion loss; ignoring directivity limits which set the floor for measurement accuracy in high-return-loss systems. Final Answer: directional coupler.

Question 9

SWR → ∞ conditions on a transmission line For a lossless line with characteristic impedance Z0, the standing-wave ratio (SWR) becomes infinite when |Γ| = 1. Which load terminations guarantee |Γ| = 1?
Select the correct combination:

Accepted Answer

Introduction / Context: SWR quantifies how severe standing waves are due to reflection from a load. Infinite SWR corresponds to complete reflection (|Γ| = 1). This occurs for specific classes of terminations and has direct implications for power handling and node voltages on lines. Given Data / Assumptions: Lossless line with characteristic impedance Z0 (real > 0). Loads considered: (1) short circuit, (2) arbitrary complex impedance, (3) open circuit, (4) purely reactive load jX. Standard definitions: Γ = (ZL − Z0) / (ZL + Z0), VSWR = (1 + |Γ|) / (1 − |Γ|). Concept / Approach: |Γ| = 1 whenever the load is lossless and not equal to Z0 in magnitude/phase such that the denominator magnitude equals the numerator magnitude. Shorts (ZL = 0) and opens (ZL → ∞) obviously yield |Γ| = 1. Purely reactive loads ZL = jX also give |Γ| = 1 because |jX − Z0| = |jX + Z0| for real Z0, leading to unit magnitude reflection (phase depends on X). A general “complex impedance” may have |Γ| < 1 if it possesses a resistive part not equal to 0 and not infinite; therefore it does not guarantee |Γ| = 1. Step-by-Step Solution: 1) Short: Γ = (0 − Z0)/(0 + Z0) = −1 → |Γ| = 1.2) Open: Γ → +1 → |Γ| = 1.3) Pure reactance: Γ = (jX − Z0)/(jX + Z0) → |Γ| = 1 (magnitudes equal). Verification / Alternative check: Compute VSWR from |Γ|; |Γ| = 1 implies denominator 0 → SWR → ∞, matching intuition of complete reflection. Why Other Options Are Wrong: Any option including “complex impedance” as a guarantee is overbroad; many complex impedances (with resistive parts) yield |Γ| < 1. Options excluding short/open/pure reactance miss guaranteed |Γ| = 1 cases. Common Pitfalls: Believing any non-matched impedance leads to |Γ| = 1; in reality, |Γ| spans (0, 1] depending on mismatch severity. Final Answer: 1, 3, and 4 (short circuit; open circuit; pure reactance).

Question 10

Waveguides vs. Coaxial Lines: Identify the True Advantages  Compared to coaxial transmission lines operating at microwave frequencies, which advantages are correctly attributed to metallic waveguides?

Accepted Answer

Introduction / Context: At microwave and millimeter-wave frequencies, engineers often choose between coaxial cables and hollow metallic waveguides. Understanding the comparative strengths of waveguides helps in high-power links, radar front-ends, and low-loss feed networks for antennas. Given Data / Assumptions: Frequency is high enough that waveguide TE/TM modes are supported (above cutoff). Both media are well-designed and properly matched to minimize reflections. Comparison focuses on intrinsic physical limits: power handling, attenuation, and practical construction. Concept / Approach: Waveguides confine fields with conducting walls supporting TE/TM modes, while coax supports a TEM mode. Surface current distribution and field strengths drive power handling and loss. Hollow guides avoid dielectric loss (usually air-filled) and distribute current over larger metal areas, usually yielding lower insertion loss and higher breakdown thresholds for a given cross-section at microwave frequencies. Step-by-Step Solution: 1) Power handling: In coax, dielectric breakdown and inner-conductor heating limit power. Waveguides, being air-filled with larger field volumes, generally withstand higher peak and average powers for comparable sizes.2) Attenuation: Coax loss includes conductor and dielectric loss. Waveguide loss is dominated by conductor loss only and often achieves substantially lower dB/m above a few GHz.3) Mechanical/Cost: Precision waveguides require machining, flanges, and alignment. They are typically more expensive and less flexible than coax, not cheaper. Verification / Alternative check: Vendor datasheets regularly show lower loss per meter and higher power ratings for standard waveguide bands (e.g., WR-xx) compared to common coax lines at the same frequency. Why Other Options Are Wrong: Option D claims waveguides are cheaper and simpler; in practice, coax is cheaper and more flexible. Option E (“all of the above”) includes D and is therefore incorrect. Options A or B alone omit the full set of true advantages, hence C is the best answer. Common Pitfalls: Assuming coax is always inferior: at lower microwave frequencies or for flexible runs, high-quality coax can be adequate. Conversely, assuming waveguide bends and twists are trivial—mechanical tolerances matter greatly. Final Answer: both (a) and (b) only

Question 11

Rectangular Waveguide Cutoff Relation (Dominant TE10)  A rectangular waveguide has broad-wall width a. At cutoff for the dominant TE10 mode, which relation between a and the cutoff wavelength λc is correct?

Accepted Answer

Introduction / Context: Cutoff conditions in rectangular waveguides determine the lowest frequency that can propagate for a given mode. For the dominant TE10 mode, the broad dimension a controls the cutoff wavelength and hence the usable frequency band for single-mode operation. Given Data / Assumptions: Uniform, air-filled rectangular waveguide with width a (broad wall) and height b (narrow wall). Dominant mode is TE10, with one half-sine variation across a and none across b. Perfect electric conductor approximation for walls. Concept / Approach: The cutoff wavelength for mode TEmn in a rectangular guide is λc = 2 / sqrt((m/a)^2 + (n/b)^2). For TE10, m = 1 and n = 0, giving λc = 2a. Rearranging yields a = λc / 2. This simple proportionality is a foundational design rule for waveguides and is frequently used for selecting a standard WR size given a desired band. Step-by-Step Solution: 1) Start from general formula: λc = 2 / sqrt((1/a)^2 + 0) = 2a.2) Rearrange: a = λc / 2.3) Therefore, at TE10 cutoff, the broad-wall width equals half of the cutoff wavelength. Verification / Alternative check: The well-known dominant-mode cutoff frequency is fc10 = c / (2a). Converting to wavelength (λc = c / fc10) again gives λc = 2a, consistent with the above. Why Other Options Are Wrong: a = λc or a = 2λc contradict the derived relation; a = λc / 4 is also incorrect; “independent of λc” ignores the defining cutoff equation. Common Pitfalls: Confusing λ (operating wavelength) with λc (cutoff wavelength); λ must be shorter than λc for propagation in a given mode. Final Answer: a = λc / 2

Question 12

Where Is Fading Most Severe? Horizon, Shadow Zone, and Good Optical Paths  Consider the statements:  Fading of signals is generally greatest near the radio horizon.  Fading of signals is generally greatest in the shadow zone.  Fading is small when a…

Accepted Answer

Introduction / Context: In terrestrial and over-the-horizon links, fading arises from multipath, diffraction, and atmospheric refractivity changes. The severity of fading depends strongly on geometry: whether the receiver is near the radio horizon, in a diffraction shadow, or has a clear line-of-sight optical path. Given Data / Assumptions: Standard atmosphere with typical refractivity gradients. Frequency in VHF/UHF/microwave where multipath and ducting can occur. “Good optical path” implies clear, unobstructed line-of-sight with strong direct component. Concept / Approach: Near the radio horizon, the direct ray skims the Earth and becomes comparable in strength to refracted or reflected rays, increasing the chance of deep interference fades. In shadow zones, the signal relies on diffraction or scattered components that are highly variable, so fading can be severe. Conversely, a high-clearance, good optical path yields a dominant direct ray and weaker multipath, so fading depth is reduced. Step-by-Step Solution: 1) Radio horizon region → competing components (direct, refracted, ground-reflected) → high fade probability.2) Shadow zone → diffractive/scattered energy only → inherently weak and variable → strong fading.3) Good optical path → strong direct LOS dominates → small fades. Verification / Alternative check: Link planning tools and ITU-R recommendations show larger fade margins required for paths near the horizon or with obstacles, and smaller margins for clear, elevated LOS paths. Why Other Options Are Wrong: Any subset that omits one of the three truths fails to capture the complete behavior across these scenarios. Common Pitfalls: Equating “radio horizon” with no signal; in practice, ducting or reflection may intermittently enhance or depress signals, producing strong fades. Final Answer: 1, 2 and 3

Question 13

Reactive Behavior of a Short, Open-Circuited Line  A transmission line segment of length less than λ/4 that is open-circuited at the far end behaves equivalently as which lumped element at the input?

Accepted Answer

Introduction / Context: Short transmission-line stubs are widely used to synthesize reactive elements for matching. The sign of the reactance seen at the input depends on whether the stub is short- or open-circuited and on its electrical length relative to λ/4. Given Data / Assumptions: Lossless line with characteristic impedance Z0. Open-circuited at the far end. Physical length ℓ is small (ℓ < λ/4) at the operating frequency. Concept / Approach: The input reactance of an open-circuited lossless line is X_in = − Z0 * cot(βℓ), where β = 2π/λ. For small electrical lengths (βℓ ≪ π/2), cot(βℓ) is large and positive, so X_in is negative (capacitive). Thus, a short open stub emulates a capacitor; by contrast, a short short-circuited stub (Z_in = j Z0 tan βℓ) is inductive for small βℓ. Step-by-Step Solution: 1) Use open-stub formula: Z_in = −j Z0 cot(βℓ).2) For 0 < βℓ < π/2, cot(βℓ) > 0 ⇒ Z_in = −j X with X > 0 (capacitive).3) Therefore, the input behaves as a capacitance. Verification / Alternative check: Smith chart movement from an open at the right edge toward the generator by a small fraction of λ rotates downward along the susceptance circle, confirming capacitive susceptance. Why Other Options Are Wrong: Inductance (A) corresponds to a short-circuited short stub; series/parallel L–C models (C, D) do not represent the pure reactance of a very short, lossless open stub; a pure resistor (E) is incorrect except at special lengths with loss. Common Pitfalls: Confusing λ/4 transformations (where opens convert to shorts) with sub-quarter-wave behavior; forgetting sign conventions for reactive input impedance. Final Answer: capacitance C

Question 14

Ridge Waveguide Interpretation  A ridge waveguide can be viewed, in first order, as which kind of modified waveguide?

Accepted Answer

Introduction / Context: Ridge waveguides introduce one or more conductive ridges into a rectangular waveguide. This structural change alters the field distribution and cutoff frequency, enabling broader bandwidth or smaller cross-sections for a given operating band. Given Data / Assumptions: Air-filled metallic waveguide with one or two ridges attached to the broad wall(s). Dominant mode under consideration (usually TE10). Losses are not the main focus here; the qualitative loading effect is. Concept / Approach: The ridges effectively increase the electric-field concentration between the ridge tips and the opposing wall. This action increases the effective capacitance between the broad walls, lowering the dominant-mode cutoff frequency. Describing the structure as “capacitively loaded” captures this first-order effect on the modal dispersion. Step-by-Step Solution: 1) Add ridges → reduce the effective gap for E-field → increase effective capacitance.2) Increased capacitance → lower TE10 cutoff → wider single-mode bandwidth for a fixed outer size.3) Hence, a ridge guide is essentially a capacitively loaded rectangular guide. Verification / Alternative check: Cutoff formulas and mode charts for ridge guides show reduced cutoff compared to smooth guides of the same external dimensions, confirming the capacitive-loading viewpoint. Why Other Options Are Wrong: Inductive loading (A) would require inserts that increase magnetic path rather than electric-field concentration; (C) refers to dielectric loading, a different technique; (D) and (E) are irrelevant to standard microwave guides. Common Pitfalls: Equating “lower cutoff” with lower loss; ridge guides usually have higher conductor loss near ridge edges. Final Answer: capacitively loaded guide

Question 15

Hybrid Tee (Magic-T) with Matched Ports  In the ideal hybrid tee shown (all ports matched): if 100 mW is fed into port 1 (an input arm), how much power is reflected back into port 1?

Accepted Answer

Introduction / Context: The hybrid tee (magic-T) is a 4-port waveguide component combining an E-plane tee and an H-plane tee. It splits and combines power with specific phase relations and, when properly matched, provides isolation between certain ports. Given Data / Assumptions: All four ports are perfectly matched (no reflections at terminations). Port 1 is used as the input (either E-arm or H-arm, depending on the convention of the figure). Ideal, lossless device. Concept / Approach: In an ideal matched hybrid tee, S11 = 0 at each port, meaning there is no reflection back into the excited port. Power applied to an E- or H-arm splits equally between the two collinear arms (with 180° relative phase for the E-arm and 0° for the H-arm) and does not appear at the isolated complementary arm. Therefore, the power reflected back into the source at the driven port is zero. Step-by-Step Solution: 1) Use the magic-T scattering matrix: S11 = 0 for a matched device.2) With an input at port 1, the device divides power to the appropriate output ports; none returns to port 1.3) Hence, reflected power at port 1 is 0 mW, independent of the input level (in the ideal case). Verification / Alternative check: Laboratory measurements of a well-matched hybrid tee show deep return loss at each port (e.g., > 20–25 dB), approximating S11 ≈ 0, so reflected power is negligible. Why Other Options Are Wrong: Any nonzero reflected power implies mismatch or internal loss, contradicting the ideal matched assumption. Common Pitfalls: Confusing isolation between E and H arms with return loss; isolation relates to cross-coupling, while return loss concerns reflections at the excited port. Final Answer: 0

Question 16

Typical Frequency Range for Satellite Communications  Which frequency range best represents the core bands commonly used for satellite communication links?

Accepted Answer

Introduction / Context: Commercial and governmental satellite communications predominantly use microwave bands where atmospheric attenuation is manageable and antennas are reasonably sized. Recognizing the typical range guides system design choices for RF front-ends, link budgets, and spectrum planning. Given Data / Assumptions: Focus on widely deployed GEO, MEO, and LEO satellite services. Terrestrial propagation losses and rain attenuation considered. Representative, not exhaustive, frequency ranges. Concept / Approach: The principal satcom bands include L band (around 1–2 GHz), S band (2–4 GHz), C band (4–8 GHz), X band (7–8/8–12 GHz), Ku band (12–18 GHz), and Ka band (26–40 GHz). The densest activity lies roughly between 1 GHz and 30 GHz, balancing antenna size, bandwidth, and atmospheric effects. Much below 1 GHz, bandwidths are small and allocations are limited; far above 30 GHz, attenuation due to rain and gases grows rapidly (though modern systems increasingly use Q/V bands with careful fade mitigation). Step-by-Step Solution: 1) Identify classic satcom bands: L, S, C, X, Ku, Ka.2) Map these to the numeric span: approximately 1–30 GHz.3) Conclude that 1 GHz to 30 GHz is the best description among the listed options. Verification / Alternative check: Regulatory allocations (e.g., ITU) and operator band plans concentrate in these ranges for uplinks and downlinks. Why Other Options Are Wrong: kHz/MHz ranges cannot carry high-throughput satellite links with practical antennas; sub-THz ranges are emerging but not the core mainstream today. Common Pitfalls: Confusing radar bands with satcom bands or assuming extremely high frequencies are the current norm for all services. Final Answer: 1 GHz to 30 GHz

Question 17

Fields at a Perfect Conductor Boundary (Normal Incidence)  A TEM plane wave is normally incident on a perfect electric conductor (PEC). At the PEC surface, what are the relative magnitudes of the electric field (E) and magnetic field (H)?

Accepted Answer

Introduction / Context: Boundary conditions at perfect conductors are a cornerstone of electromagnetic theory. They determine reflection behavior and field distributions in waveguides, cavities, and scattering problems. Given Data / Assumptions: Plane TEM wave at normal incidence. Perfect electric conductor (infinite conductivity) so tangential E at the surface must vanish. Time-harmonic steady state. Concept / Approach: At a PEC boundary, the tangential component of the electric field is zero. The incident and reflected waves superpose to enforce E_t = 0 at the surface, producing a pressure node (minimum). The magnetic field forms a standing-wave antinode (maximum) at the surface, driving surface currents necessary to cancel the tangential E. Step-by-Step Solution: 1) Apply boundary condition: E_t(surface) = 0 for a PEC.2) Incident + reflected fields sum to zero tangential E at z = 0 → E minimum (zero).3) H_t(surface) doubles relative to a single traveling wave due to constructive interference → H maximum. Verification / Alternative check: Using standing-wave relations for normal incidence, the E field has a node at the surface; the H field has an antinode at the surface, consistent with power not penetrating the PEC. Why Other Options Are Wrong: Options B, C, and D contradict the PEC boundary conditions; E cannot be maximum at the surface, and H need not vanish. Common Pitfalls: Confusing E and H boundary behavior; assuming both go to zero at metal surfaces, which is false for H except in special cases inside the conductor with nonzero loss. Final Answer: minimum and maximum

Question 18

LED vs. LASER: Which Properties Are Superior for a LASER Source?  Consider the statements comparing a LASER source with an LED:  Higher emission efficiency.  No tuning arrangement.  Narrow spectral width.  Provision for optical confinement. Which st…

Accepted Answer

Introduction / Context: Semiconductor LEDs and LASER diodes are both widely used optical sources. However, they differ fundamentally in emission mechanism and optical characteristics, which drives their use in communications, sensing, and illumination. Given Data / Assumptions: Typical edge-emitting or VCSEL-type LASER diodes versus common LEDs. Comparison at similar emission wavelengths. General properties rather than specific niche devices. Concept / Approach: LASERs provide stimulated emission with optical feedback, producing a coherent, narrow spectral line (small Δλ) and strong optical confinement in a cavity and waveguide. LEDs emit by spontaneous recombination over a broader spectrum and generally lower external efficiency into a given optical mode. Step-by-Step Solution: 1) Efficiency: Many LASER diodes exhibit higher slope and wall-plug efficiencies into single-mode fibers than comparable LEDs → statement 1 true in typical communications contexts.2) Tuning: LASERs can be tuned via temperature, current, or external cavities; saying “no tuning arrangement” is incorrect → statement 2 false.3) Spectral width: LASERs have narrow linewidth; LEDs are broadband → statement 3 true.4) Confinement: LASERs use optical cavities and waveguiding for mode control → statement 4 true. Verification / Alternative check: Datasheets and textbooks show LED spectral widths of tens of nanometers versus LASER linewidths from sub-nanometer to much narrower; also, coherent beam quality and coupling efficiencies are higher for LASERs. Why Other Options Are Wrong: Any option including statement 2 is invalid because LASERs indeed can be tuned or include provision for tuning. Common Pitfalls: Assuming “efficiency” only means radiant flux per electrical watt without considering coupling into an optical mode; LASERs excel at delivering power into single-mode fibers. Final Answer: 1, 3, 4 are correct

Question 19

Magic Tee (Hybrid Tee) Port Isolation Properties  Consider the following statements about a magic tee:  The two collinear arms are isolated from each other.  One of the collinear arms is isolated from the E arm.  One of the collinear arms is isolate…

Accepted Answer

Introduction / Context: The magic tee combines E-plane and H-plane tees to form a 4-port microwave junction used for splitting/combining with specific phase characteristics and port isolations. Knowing which ports are isolated is essential for correct network design. Given Data / Assumptions: Ports: two collinear arms and two side arms (E arm and H arm). Ideal, lossless, perfectly matched device (Sii = 0). Standard port numbering where the side arms correspond to E and H excitations. Concept / Approach: In the ideal magic tee, the collinear arms do not couple directly to each other (S12 = 0), giving isolation between them. Further, the E and H arms are isolated (S_EH = 0), so a signal injected into one does not appear at the other. However, each collinear arm couples to both E and H arms with specific phase relations, so no single collinear arm is isolated from either side arm on its own. Step-by-Step Solution: 1) Use the magic-tee S-matrix: S12 = 0 → collinear arms isolated → statement 1 true.2) S_EH = 0 → E and H arms isolated → statement 4 true.3) Coupling from E or H arms splits equally to both collinear arms → statements 2 and 3 false. Verification / Alternative check: Standard references show equal-magnitude in-phase (H-arm) or out-of-phase (E-arm) splits to the collinear ports, with isolation between the collinear pair and between the E and H arms. Why Other Options Are Wrong: Options that include 2 or 3 assert an isolation that does not exist; energy from E or H arms does reach both collinear arms. Common Pitfalls: Confusing “isolation” with “no net output due to phasing”; even when outputs cancel at a combining node, the ports themselves are not isolated unless the S-parameter is zero. Final Answer: 1 and 4 are correct

Question 20

Directional coupler — primary purpose in microwave measurements In RF and microwave engineering, a directional coupler is used for which of the following tasks?

Accepted Answer

Introduction / Context: A directional coupler is a four-port passive device that samples a defined fraction of power travelling in a transmission line, with strong directivity so that forward and reverse waves can be observed separately. Engineers rely on it to characterize systems (VSWR, return loss, gain, isolation) without interrupting the main signal path. Given Data / Assumptions: Ideal or well-designed directional coupler with specified coupling (for example, 10 dB, 20 dB) and high directivity. Coupler placed inline with a source and load to pick off samples of forward and reflected waves. Appropriate detectors or a vector network analyzer to read amplitude and phase. Concept / Approach: The coupler provides two auxiliary ports: a “coupled” port that samples the forward-travelling wave and an “isolated” port that ideally rejects it (and vice versa for the reverse direction). By measuring voltages at the sampled ports with phase-sensitive instruments, one can determine the magnitude and phase of travelling waves, enabling calculation of reflection coefficient, VSWR, and S-parameters. Step-by-Step Solution: Insert the coupler in the main line between source and load.Connect power sensors or a VNA receiver to the coupled port(s) to read forward and reverse samples.From measured samples, compute |Γ|, phase(Γ), and derived quantities such as return loss and VSWR. Verification / Alternative check: With a known calibration standard (open/short/match), readings at the coupler ports replicate expected Γ magnitudes and phases, verifying the device’s intended measurement role. Why Other Options Are Wrong: Transmit microwave signals: transmission occurs via the main line; the coupler is a sampler, not a transmitter. Generate microwave signals: generation is the role of oscillators, klystrons, Gunn sources, etc. Both (a) and (b): conflates measurement hardware with sources/lines. Common Pitfalls: Confusing coupling (fixed fraction) with insertion loss; ignoring directivity limits which affect accuracy of reflected-wave measurement; using the wrong coupling value, leading to either overload or too little signal at the detector. Final Answer: to measure amplitude and phase of a travelling wave

Microwave Communication Questions