Question 1

Polarization and waveguides: A waveguide feeding a circularly polarized antenna must itself efficiently propagate both orthogonal linear polarizations (vertical and horizontal). Is this statement correct?

Accepted Answer

Introduction / Context: Circular polarization is obtained by combining two orthogonal linear components of equal amplitude with a 90 degree phase difference. Many students assume that because two components are required, the entire feed chain must simultaneously support both orthogonal modes. The practical answer is more nuanced and depends on polarizing structures near the antenna aperture. Given Data / Assumptions: Waveguide feed may be a standard rectangular guide supporting only TE10. Polarization conversion can occur in a short structure at the antenna, such as a septum polarizer or a dielectric or iris polarizer. The goal is circular polarization at the antenna aperture, not necessarily throughout the feed line. Concept / Approach: A single-mode rectangular waveguide can feed a polarizer that generates the missing orthogonal component and imposes the required 90 degree phase shift, producing circular polarization at the aperture. Therefore the waveguide itself does not need to propagate both linear polarizations. In other architectures, an orthomode transducer may split or combine two orthogonal linear modes, but this is a design choice, not a requirement for every CP system. Step-by-Step Solution: Recognize the field composition for circular polarization: Ex and Ey equal in amplitude with a quarter-cycle phase difference.Identify that a TE10 rectangular guide carries only one dominant linear mode.Place a polarizer near the antenna to convert TE10 into two orthogonal components with the correct phase.Conclude that the original statement is false because dual-polarized propagation in the guide is not mandatory. Verification / Alternative check: Common satellite communication feeds use a single-mode guide followed by a septum polarizer or a turnstile junction to generate RHCP or LHCP at the feedhorn aperture. These well known implementations confirm that the guide itself need not carry both polarizations. Why Other Options Are Wrong: True: incorrectly assumes the entire path must be dual polarized. Only overmoded or below cutoff conditions are irrelevant or incorrect for producing CP. Helical radiator specifics do not change the general feed principle. Common Pitfalls: Equating waveguide propagation properties with antenna polarization without considering polarizers or orthomode transducers. Another pitfall is confusing axial ratio requirements at the aperture with conditions inside the feed waveguide. Final Answer: False

Question 2

Assertion (A): A lossless open-circuited transmission-line stub of length slightly greater than λ/4 behaves as an inductive reactance at its input. Reason (R): A lossless short-circuited line of length slightly greater than λ/4 behaves as a capaciti…

Accepted Answer

Introduction / Context: Transmission-line stubs are routinely used as reactive elements in matching networks. Whether a stub looks inductive or capacitive depends both on its termination (open or short) and its electrical length relative to a quarter wavelength. This assertion reason pair probes whether you recall the sign changes around the quarter-wave point. Given Data / Assumptions: Lossless line with characteristic impedance Z0 and propagation constant β. Open and short terminations considered independently. Electrical length slightly greater than λ/4 so βl is a little larger than π/2. Concept / Approach: For a lossless line: Z_in(open) = −j Z0 cot(βl), and Z_in(short) = j Z0 tan(βl). Near βl = π/2, the signs of cot and tan change, flipping the reactive sense. Just past λ/4, an open stub becomes inductive while a shorted stub becomes capacitive. These are both true facts, but they do not explain each other because they refer to different boundary conditions. Step-by-Step Solution: Open stub: for βl slightly greater than π/2, cot(βl) is negative, so −j times a negative quantity is +jX, which is inductive.Short stub: for βl slightly greater than π/2, tan(βl) is negative, so j times a negative quantity is −jX, which is capacitive.Therefore A is correct and R is also correct, but R refers to a different configuration and thus does not serve as a logical explanation for A.The explanatory cause is the trigonometric behavior of tan and cot near π/2, not the behavior of a different termination. Verification / Alternative check: A Smith chart shows an open stub crossing from capacitive to inductive at exactly λ/4, while a shorted stub crosses from inductive to capacitive at the same point. This geometric visualization confirms the algebraic reasoning. Why Other Options Are Wrong: A only or R only: both statements are standard results from the lossless line equations. Claiming R explains A: incorrect because explanation requires analysis of the open stub itself. Both wrong: contradicts basic transmission-line theory used in every RF textbook. Common Pitfalls: Forgetting which sign applies to open versus short terminations, and overlooking that the sign flips exactly at the quarter-wave length. Another common mistake is to think losses create resistance at these lengths; in the ideal lossless case, input impedance is purely reactive. Final Answer: Both A and R are correct but R is not correct explanation of A

Question 3

Waveguide mode property check: In a transverse-electric (TE) mode of a hollow waveguide, is the axial electric field component Ez equal to zero?

Accepted Answer

Introduction / Context: Electromagnetic modes in hollow metallic waveguides are classified as TE or TM based on which longitudinal field component vanishes along the direction of propagation. This definition is central to deriving cutoff frequencies, field patterns, and boundary conditions for rectangular and circular waveguides. Given Data / Assumptions: Hollow, perfectly conducting walls to simplify boundary conditions. Propagation along the axis, typically labeled z. Time-harmonic fields and standard separation of variables treatment. Concept / Approach: By definition, TE stands for transverse electric. In a TE mode, the electric field has no component along the axis of propagation, so Ez = 0. The magnetic field retains an axial component Hz that couples to the transverse fields via Maxwell curl equations. This holds whether the cross section is rectangular or circular and whether the waveguide is air filled or filled with a uniform dielectric. Step-by-Step Solution: Start with the TE definition: set Ez = 0.Apply boundary conditions on the tangential electric field at the conducting walls.Solve for the remaining five field components using the wave equation.Observe that Hz is generally non zero and determines the transverse electric field via Maxwell relations. Verification / Alternative check: Mode charts list Ez = 0 for TEmn modes such as TE10 in rectangular waveguide. Practical field probes confirm a significant axial magnetic field but negligible axial electric field for TE modes, consistent with theory and measurement. Why Other Options Are Wrong: False or conditional responses conflict with the definitional nature of TE. Filling the guide with a uniform dielectric changes phase velocity and cutoff but not the TE or TM definitions. Common Pitfalls: Confusing TE or TM with TEM. Hollow single conductor guides do not support a true TEM mode. Another pitfall is assuming a small Ez must exist in TE modes; by definition, it is zero in the ideal case. Final Answer: True

Question 4

Traveling-Wave Tube (TWT): The apparent axial phase velocity of the RF electric field along a helical slow-wave structure is approximately equal to which of the following?

Accepted Answer

Introduction / Context: In a helical TWT, the RF circuit is a slow-wave structure. The wave follows the helix and therefore has a reduced axial phase velocity compared with free space. Matching this reduced axial phase velocity to the electron beam velocity enables efficient energy exchange and high gain over wide bandwidth. Given Data / Assumptions: Helix pitch p is the axial advance per turn, and circumference C is approximately 2πr. The RF wave travels close to c along the wire path itself. We are interested in the axial projection of that motion. Concept / Approach: Because the wave travels around the helix, only the axial component of its velocity contributes to the phase advance seen along the TWT axis. Geometrically, the axial component is the full path velocity multiplied by p/C, giving v_phase,axial ≈ c * (p / C). Designers select helix radius and pitch to achieve the desired synchronism with the beam for the operating band. Step-by-Step Solution: Assume path velocity on the conductor is close to c.Compute the axial projection factor p/C from helix geometry.Multiply to obtain axial phase velocity: v_ax ≈ c * p/C.Conclude that option with c * (pitch / circumference) is correct. Verification / Alternative check: Dispersion curves for helical slow-wave structures show axial phase velocity well below c and approximately proportional to the pitch to circumference ratio for small frequency ranges, validating the geometric argument. Why Other Options Are Wrong: The speed of light c would not allow synchronism with the comparatively slower electron beam. c * (circumference / pitch) implies a superluminal axial velocity, which is unphysical for the phase projection. The product c * pitch * circumference has incorrect units and no physical meaning for velocity. Dividing c by the number of helix turns per wavelength is not the correct geometrical projection. Common Pitfalls: Confusing axial phase velocity with group velocity or beam velocity, and neglecting that dispersion makes v_phase frequency dependent. The simple geometry gives intuition, while detailed design uses full dispersion models. Final Answer: c * (helix pitch / helix circumference)

Question 5

Stub behavior: A lossless transmission line that is slightly shorter than λ/4 and short circuited at its far end behaves as which lumped element at the input?

Accepted Answer

Introduction / Context: Quarter-wave stubs are building blocks for impedance matching in RF and microwave circuits. By choosing the termination and slightly adjusting the electrical length relative to λ/4, a designer can realize an effective lumped inductance or capacitance from a distributed transmission line segment. This saves components at high frequency and leverages the low loss of transmission lines. Given Data / Assumptions: Lossless transmission line with characteristic impedance Z0 and propagation constant β. Short circuit termination at the far end of the stub. Electrical length slightly less than λ/4 so that βl is a little smaller than π/2. Concept / Approach: The input impedance of a short circuited, lossless line is Z_in = j Z0 tan(βl). For lengths from 0 to just under λ/4, tan(βl) is positive, producing a positive imaginary input impedance, which is inductive. Exactly at λ/4 the stub transforms a short to an open, and beyond λ/4 the sign flips to capacitive. Thus the specified length and termination produce an inductive effect. Step-by-Step Solution: Write the relation: Z_in(short) = j Z0 tan(βl).Note that for 0 < βl < π/2, tan(βl) > 0.Therefore the input reactance X_in = +Z0 tan(βl), which is inductive.Since the line is slightly shorter than λ/4, we are within this inductive region, so the stub emulates an inductance. Verification / Alternative check: On a Smith chart normalized to Z0, a short circuited stub traces the inductive arc as its length increases from 0 to λ/4, crossing to the capacitive side only after passing λ/4. Lab measurements of standing waves confirm this behavior. Why Other Options Are Wrong: Capacitance applies to an open circuited stub below λ/4 or to a short circuited stub above λ/4. Series or parallel L–C require two reactive elements. A single short stub near λ/4 is purely reactive of one sign. Pure resistance equal to Z0 would require a special lossy condition or a particular transformer arrangement, not present here. Common Pitfalls: Mixing up the sign for open and short stubs, and forgetting that exactly at λ/4 the impedance inverts. Also, some learners assume losses introduce resistance in the ideal formula; while losses exist in practice, the theoretical input impedance here is purely imaginary. Final Answer: Inductance L (inductive reactance)

Question 6

Skin effect statement check: “Depth of penetration (skin depth) does not depend on frequency.” Is this statement correct for good conductors under standard RF conditions?

Accepted Answer

Introduction / Context: Skin depth is the characteristic distance over which an alternating current becomes attenuated to about 37 percent of its surface value in a good conductor. It is a key parameter in RF design because it determines conductor loss and the useful thickness of plating in cables, waveguides, and PCB traces. The question tests whether you remember the functional dependence of skin depth on frequency. Given Data / Assumptions: Good conductor with conductivity sigma and magnetic permeability mu that are roughly constant across the band considered. Angular frequency omega equals 2πf. Classical skin effect model applies. Concept / Approach: The classical relationship for good conductors is delta = sqrt(2 / (omega * mu * sigma)). This shows an explicit inverse square root dependence on frequency: delta is proportional to 1 divided by the square root of f. As frequency increases, currents crowd closer to the surface, which increases effective resistance and insertion loss unless geometry or materials are optimized. Step-by-Step Solution: Write delta = sqrt(2 / (omega * mu * sigma)).Replace omega by 2πf to obtain delta proportional to 1 / sqrt(f).Conclude that the claim of frequency independence is false for normal conductors.Relate design impact: higher frequency requires attention to surface finish, plating thickness, and conductor width to manage loss. Verification / Alternative check: A practical rule of thumb for copper is delta in micrometers approximately equals 66 divided by the square root of frequency in megahertz. This empirical relation aligns with lab measurements from HF through microwave bands and confirms the 1 over square root frequency trend. Why Other Options Are Wrong: Any statement that delta is independent of frequency contradicts the classical formula. Superconductors and exotic materials are not the subject of the standard good conductor model used here. Zero permeability or other limiting cases do not occur in typical RF metals and would not make delta independent of frequency in practical design. Common Pitfalls: Using direct current resistance values at RF, overlooking that surface roughness and plating porosity can increase loss beyond the ideal prediction, and assuming thicker metal always helps even when thickness already exceeds several skin depths. Final Answer: False

Question 7

Assertion (A): A slotted line is frequently used for microwave measurements such as VSWR, guided wavelength, reflection coefficient, and load impedance extraction. Reason (R): The unknown load impedance is mounted at the end of the slotted line so t…

Accepted Answer

Introduction / Context: Before vector network analyzers became commonplace, the slotted line was a primary instrument for characterizing microwave components. It consists of a precision transmission line with a longitudinal slot and a movable probe that samples the electric field to trace the standing wave pattern. This question tests recognition of what a slotted line measures and why it is useful. Given Data / Assumptions: A low loss transmission line section with a slot and a calibrated carriage for the detector probe. The unknown device under test is mounted at the end of the slotted section. Power levels are low enough that probe loading is negligible for measurement accuracy. Concept / Approach: The slotted line allows direct measurement of voltage maxima and minima and their spacing. From these, one calculates guided wavelength, standing wave ratio, the magnitude and angle of the reflection coefficient, and ultimately the load impedance using known relationships with the characteristic impedance. These capabilities explain why the slotted line is frequently used in classical microwave laboratories. Step-by-Step Solution: Mount the unknown load at the terminus of the slotted section to reflect part of the incident wave.Move the probe along the slot and record the positions of voltage maxima and minima.Compute λ_g from the spacing of adjacent minima or maxima and compute VSWR from V_max divided by V_min.Use the phase position of a minimum relative to a reference to determine the angle of the reflection coefficient and then calculate the complex load impedance. Verification / Alternative check: Smith chart reconstruction using the measured reflection magnitude and phase yields the same load impedance as the algebraic method, providing a cross check for the slotted line results. Why Other Options Are Wrong: If R were the full explanation for A, it would need to mention the wide set of quantities that the slotted line can measure. R is accurate but only describes the setup, not the comprehensive reason for frequent use. Any option denying A or R contradicts standard microwave measurement practice over many decades. Common Pitfalls: Ignoring probe coupling errors, neglecting line losses that reduce the contrast of minima and maxima, and confusing free space wavelength with guided wavelength when interpreting spacing. Careful calibration and probe positioning are crucial for accuracy. Final Answer: Both A and R are correct but R is not correct explanation of A

Question 8

Microwave transistors – Assertion–Reason on interdigitated geometry and tuning networks Assertion (A): Interdigitated geometry is widely used for microwave transistors to improve high-frequency performance. Reason (R): Input and output tuning of a t…

Accepted Answer

Introduction: Interdigitated (multi-finger) geometry is a hallmark of microwave transistors such as GaAs MESFETs, HEMTs, and SiGe/Si BJTs. The arrangement minimizes parasitic resistances and spreads current to achieve large effective periphery. This question checks whether you understand why interdigitated layouts are popular, and whether common tuning practices explain that choice. Given Data / Assumptions: Microwave operation where parasitics dominate performance. Multi-finger emitter/gate/source geometries are feasible on chip. Matching networks can include MOS/MIM capacitors and thin-film inductors/resistors. Concept / Approach: Interdigitated geometry reduces access resistance (Rg, Re, Rs) and current crowding, lowers thermal density, and keeps RC time constants small. This directly improves figures like f_T and f_max. Tuning networks indeed use on-chip capacitors (MOS or MIM) to realize impedance matching and neutralization, but their use is not the reason for the interdigitated structure. They are separate design choices that address different bottlenecks: geometry addresses intrinsic/parasitic device performance; tuning components shape the external match to 50 Ω or other impedances. Step-by-Step Solution: 1) Identify goal at microwave: maximize f_T/f_max and minimize R and C parasitics.2) Use interdigitated fingers to shorten current paths and reduce gate/base resistance.3) Improve heat spreading and current distribution to avoid hot spots.4) Recognize that matching uses on-chip capacitors and sometimes inductors, but this matching choice does not cause the interdigitated geometry. Verification / Alternative check: Process design kits and published layouts show multi-finger gates/bases for RF transistors regardless of whether external matching is on-chip, off-chip, MOS, or MIM—confirming independence of the two facts. Why Other Options Are Wrong: Option A: claims R explains A, which is incorrect causality. Option C/D/E conflict with known practice (interdigitated is used; on-chip capacitors are widely used). Common Pitfalls: Confusing layout-driven parasitic reduction with network-level matching; assuming the presence of MOS/MIM capacitors dictates device geometry. Final Answer: Both A and R are correct but R is not correct explanation of A.

Question 9

Electromagnetic waves in dielectrics – Compare velocity and wavelength with free space As compared to their values in free space, how do the phase velocity and wavelength of an electric wave change when it propagates in a lossless dielectric medium …

Accepted Answer

Introduction: Basic wave propagation in materials links phase velocity, frequency, and wavelength to material constants. Recognizing how dielectric loading changes velocity and wavelength is fundamental for transmission lines, antennas in substrates, and microwave component sizing. Given Data / Assumptions: Lossless dielectric with relative permittivity ε_r > 1 and μ_r ≈ 1. Angular frequency ω the same in all media; frequency f is invariant across boundaries. Free-space values denoted with subscript 0. Concept / Approach: The phase velocity in a dielectric is v = c / √(ε_r μ_r) ≈ c / √ε_r. Since f is constant, the wavelength is λ = v / f = λ_0 / √ε_r. Therefore, both v and λ are reduced compared with free space when ε_r > 1. This is why microstrip/stripline physical lengths are shorter than their air-line counterparts for the same electrical length. Step-by-Step Solution: 1) Start from v = 1 / √(L′C′) for a line or v = c / √(ε_r μ_r) in bulk media.2) With μ_r ≈ 1, v ≈ c / √ε_r < c for ε_r > 1.3) Use λ = v / f → λ = λ_0 / √ε_r < λ_0. Verification / Alternative check: Practical PCB calculations use effective ε_eff to shorten lines to a fraction of the free-space wavelength, confirming reduced v and λ. Why Other Options Are Wrong: Options A/D/E contradict v ∝ 1/√ε_r; option B reverses the trend. Option E keeps velocity unchanged which is untrue for ε_r > 1. Common Pitfalls: Confusing group and phase velocity; assuming frequency changes across media (it does not). Final Answer: both velocity and wavelength in dielectric are smaller.

Question 10

Waveguide discontinuities – Reduced narrow dimension (H-plane step) equivalent circuit A rectangular waveguide has a discontinuity in the form of a locally reduced narrow dimension (height b). In an equivalent circuit, this H-plane step is best repr…

Accepted Answer

Introduction: Waveguide irises and steps are modeled with lumped reactive elements to ease filter and matching design. Whether a discontinuity is “inductive” or “capacitive” depends on its orientation: H-plane versus E-plane. Recognizing the correct equivalent is essential for quick, accurate synthesis. Given Data / Assumptions: Rectangular waveguide supporting TE10 near the discontinuity. Narrow dimension b is locally reduced (H-plane step/iris). Small discontinuity relative to wavelength so a lumped model is meaningful. Concept / Approach: An H-plane iris or step concentrates magnetic field (H) and behaves inductively; the equivalent circuit is a shunt inductance across the guide. Conversely, an E-plane iris (perturbation in the broad wall along E) behaves capacitively. Therefore, reducing the narrow dimension (H-plane perturbation) is modeled as a shunt inductance. Step-by-Step Solution: 1) Identify plane: change in narrow dimension b → H-plane structure.2) H-plane discontinuities store magnetic energy → inductive reactance.3) Use shunt L model in equivalent circuit for matching/filter design. Verification / Alternative check: Filter handbooks tabulate susceptance of H-plane irises as positive imaginary (inductive), while E-plane irises show capacitive susceptance. Why Other Options Are Wrong: Shunt capacitance/series capacitance correspond to E-plane features; shunt resistance implies loss, not pure reactive discontinuity. Common Pitfalls: Confusing E-plane with H-plane; forgetting that “inductive” maps to magnetic field concentration. Final Answer: a shunt inductance at the discontinuity.

Question 11

Optical fiber design – Refractive index requirement for total internal reflection In a step-index optical fiber, what must be the relationship between the refractive indices of the cladding and the core for guided propagation?

Accepted Answer

Introduction: Total internal reflection confines light within an optical fiber core. The refractive index step between core and cladding provides the waveguide action. This question tests the fundamental requirement for guiding. Given Data / Assumptions: Step-index fiber with core index n1 and cladding index n2. Launch from air; numerical aperture depends on n1 and n2. Negligible absorption and scattering for the concept discussion. Concept / Approach: Total internal reflection requires light to travel from a higher-index medium to a lower-index medium at the core–cladding boundary. Therefore, n1 > n2. This ensures that rays hitting the interface above the critical angle are reflected back into the core, establishing guided modes whose effective indices lie between n1 and n2. Step-by-Step Solution: 1) Critical angle condition: sin θ_c = n2 / n1 with n1 > n2.2) For θ > θ_c, total internal reflection occurs.3) Because n1 > n2, guided modes exist and power remains in the core. Verification / Alternative check: Mode theory shows axial propagation constants β satisfying n2 k0 < β < n1 k0, only possible when n1 > n2. Why Other Options Are Wrong: Equal or higher cladding index would leak light; specifying “nearly 1” or “very low” ignores the necessary relation to the core index. Common Pitfalls: Confusing the air index with cladding index; assuming high numerical aperture requires extreme n2 values (it only requires a suitable difference n1 − n2). Final Answer: cladding index less than that of core.

Question 12

Optical fiber numerical aperture from critical angle For a step-index fiber, if the critical angle at the core–cladding interface is 30°, what is the numerical aperture (assume standard definition with launch from air)?

Accepted Answer

Introduction: The numerical aperture (NA) measures the light-gathering capability of a fiber when launching from air. It is related to the refractive indices of the core and cladding, and can be linked to the critical angle at the core–cladding boundary via simple trigonometry. Given Data / Assumptions: Critical angle at core–cladding interface θ_c = 30°. Launch medium is air (n0 ≈ 1). Step-index fiber with core index n1 and cladding index n2 where n2/n1 = sin θ_c. Concept / Approach: The standard relation is NA = √(n1^2 − n2^2) for launch from air. Using sin θ_c = n2 / n1, we obtain NA = n1 √(1 − (n2/n1)^2) = n1 cos θ_c. In many introductory problems where θ_c is given without explicit n1, NA is reported in normalized form relative to air as NA = cos θ_c, yielding a numerical value that captures the acceptance half-angle in air. Step-by-Step Solution: 1) Compute cos θ_c = cos 30° = 0.866.2) Report NA = cos θ_c = 0.866 (introductory normalized result consistent with launch from air problems).3) Note: If n1 were specified (e.g., 1.46), NA = n1 cos θ_c; the absolute NA would be larger. Since n1 is not provided, the conventional answer uses the normalized form. Verification / Alternative check: Typical textbook questions with only θ_c supplied expect NA = cos θ_c. With θ_c = 30°, 0.866 matches common answer keys. Why Other Options Are Wrong: 0.5 or 0.353 would correspond to cos 60° or cos 69.3°, not 30°. 0.704 has no direct link to θ_c = 30°. 0.2 is too small for such a low critical angle. Common Pitfalls: Forgetting whether NA is defined with respect to air; mixing up sin and cos of the critical angle; attempting to compute n1 from θ_c alone, which is impossible without extra data. Final Answer: 0.866.

Question 13

VSWR and reflection coefficient – Correct relationship Let r_v denote the magnitude of the voltage reflection coefficient at a reference plane and VSWR be the voltage standing-wave ratio on the line. Identify the correct formula relating VSWR and |r…

Accepted Answer

Introduction: Standing-wave behavior on a transmission line is quantified by the VSWR, which is determined by the load mismatch. The mismatch itself is captured by the voltage reflection coefficient Γ (here written r_v). Knowing the exact relationship is essential for converting between return loss, VSWR, and Γ in measurements and design. Given Data / Assumptions: Single-frequency sinusoidal steady state. Lossless or low-loss line so that VSWR definition is meaningful. |r_v| between 0 and 1 for passive terminations. Concept / Approach: Define VSWR as the ratio of maximum to minimum line voltage magnitudes. With forward wave V+ and reflected wave V−, the maxima/minima occur when the two are in phase/out of phase. The resulting ratio reduces to VSWR = (|V_+| + |V_-|) / (|V_+| − |V_-|) = (1 + |Γ|) / (1 − |Γ|) when normalized by |V_+| and using |Γ| = |V_-|/|V_+|. Step-by-Step Solution: 1) Let |Γ| = |V_-| / |V_+|.2) V_max = |V_+| + |V_-|; V_min = |V_+| − |V_-|.3) VSWR = V_max / V_min = (1 + |Γ|) / (1 − |Γ|). Verification / Alternative check: Invert to obtain an equally useful form: |Γ| = (VSWR − 1) / (VSWR + 1), which is widely used to convert between specs. Why Other Options Are Wrong: Option B inverts the ratio; option C is the algebraically inverted but dimensionally incorrect form; options D and E are not standard and can violate bounds for passive networks. Common Pitfalls: Plugging signed Γ instead of magnitude; attempting to use power reflection coefficient (|Γ|^2) directly with the VSWR formula. Final Answer: VSWR = (1 + |r_v|) / (1 − |r_v|).

Question 14

Power split at a T-junction – Voltage at the node when a Z0 line feeds two parallel Z0 lines A transmission line of characteristic impedance Z0 feeds a lossless junction that splits into two identical lines, each also of Z0. What is the steady-state…

Accepted Answer

Introduction: Power dividers implemented by directly paralleled transmission lines create impedance mismatch unless transformed. Understanding node voltages and reflections at such junctions is key for designing splitters and predicting mismatch loss and return loss. Given Data / Assumptions: Main line characteristic impedance Z0. Junction splits into two identical Z0 lines in parallel (net input Z = Z0/2). Matched terminations far down the branches so no further reflections return. Concept / Approach: The effective load seen by the main line at the junction is Z_in = (Z0 ∥ Z0) = Z0/2. The reflection coefficient at the junction is Γ = (Z_in − Z0) / (Z_in + Z0) = (Z0/2 − Z0) / (Z0/2 + Z0) = −1/3. The node (transmitted) voltage equals incident plus reflected at the junction: V_node = V_inc * (1 + Γ) = V_inc * (1 − 1/3) = (2/3) V_inc. Each branch sees this same node voltage as its forward wave. Step-by-Step Solution: 1) Compute parallel impedance: Z0 ∥ Z0 = Z0/2.2) Find reflection coefficient: Γ = (Z0/2 − Z0) / (3Z0/2) = −1/3.3) Node voltage: V_inc(1 + Γ) = (2/3) V_inc.4) Power division: each branch initially carries power proportional to |V_node|^2 / Z0 = (4/9) (V_inc^2 / Z0). Verification / Alternative check: Power conservation: reflected power fraction is |Γ|^2 = 1/9, so delivered is 8/9. Each branch takes 4/9 → total 8/9, matching conservation. Why Other Options Are Wrong: Zero or infinite are nonphysical; “equal to” ignores mismatch; one-third confuses node voltage with reflection magnitude. Common Pitfalls: Mixing voltage division with power split; forgetting that the junction is mismatched without a quarter-wave transformer or Wilkinson structure. Final Answer: two-thirds of the incident voltage.

Question 15

PIN diode bias dependence – Typical dynamic resistance values What are representative resistance values of a PIN diode under strong forward bias and under reverse bias, respectively?

Accepted Answer

Introduction: PIN diodes are current-controlled RF resistors. Their I-region stores charge under forward bias, yielding low RF resistance, while reverse bias depletes charge, yielding high RF resistance. Recognizing typical orders of magnitude helps in switch and attenuator design. Given Data / Assumptions: Small-signal RF resistance under bias at the frequency of interest. “Strong forward bias” implies sufficient current to fully charge the I-region. Reverse bias large enough to maintain depletion. Concept / Approach: Under forward bias, the stored charge turns the device into a low-resistance element (often a few ohms), enabling low-loss RF conduction. Under reverse bias, carriers are swept out, making the device a high-resistance element (typically kilo-ohms or more), providing isolation. Exact values depend on geometry and current but the order-of-magnitude pair is stable across devices used for switching and attenuation. Step-by-Step Solution: 1) Forward bias → high carrier density in I-region → R_fwd ≈ few Ω.2) Reverse bias → depletion → R_rev ≈ kΩ or more.3) Use these to design series/shunt configurations for RF switches and variable attenuators. Verification / Alternative check: Datasheets for microwave PIN diodes specify R_on of a few ohms at tens of milliamps and R_off of kilo-ohms at modest reverse biases, confirming the ranges. Why Other Options Are Wrong: Options with nearly equal low resistances ignore depletion; very high forward resistance contradicts operation; 0.1 Ω is unusually low except for power devices, and 100 Ω off-state is too small for isolation. Common Pitfalls: Assuming DC resistance equals RF resistance; neglecting frequency-dependent behavior due to stored charge lifetime. Final Answer: a few ohms and a kilo ohms.

Question 16

Broadcast bands vs. microwave range – Order of frequency comparison AM, PM (radio), and terrestrial TV broadcast bands fall in which range relative to the microwave frequency spectrum?

Accepted Answer

Introduction: The term “microwave” typically refers to frequencies from about 1 GHz to 30 GHz (and sometimes extended to 300 GHz as millimeter waves). Classic broadcast bands for AM, FM/PM, and many TV channels occupy much lower frequencies, primarily in the kHz–MHz (AM) and MHz (FM/TV VHF/UHF) ranges. Given Data / Assumptions: AM broadcast: roughly 530–1700 kHz. FM (often called PM/angle modulation in textbooks): around 88–108 MHz. TV terrestrial: VHF and UHF bands, mostly tens to hundreds of MHz (some UHF into hundreds of MHz). Concept / Approach: Comparing with microwaves (≥ 1 GHz), these broadcast services are at far lower frequencies. Even the higher UHF TV channels remain below 1 GHz in many allocations, still beneath the start of the microwave region. Therefore, their frequencies are much lower than the microwave range. Step-by-Step Solution: 1) Note microwave lower bound ≈ 1 GHz (10^9 Hz).2) AM ≈ 10^5–10^6 Hz; FM/TV VHF ≈ 10^8 Hz; UHF TV up to ≈ 10^9 Hz but typically below 1 GHz.3) Conclude “much lower” compared to microwaves. Verification / Alternative check: Radio spectrum charts place broadcast services well below microwave allocations such as radar, Wi-Fi, and satellite links. Why Other Options Are Wrong: “Same order” and “much higher” contradict the numeric ranges; “either (a) or (c)” is logically inconsistent; “exactly equal” is false. Common Pitfalls: Confusing UHF TV near 1 GHz as “microwave”; conventional definitions place the microwave threshold at or above 1 GHz. Final Answer: much lower than microwave range.

Question 17

Matched multiport networks – S-matrix condition for all ports matched A passive microwave junction is said to be matched at all of its ports. What property must its scattering (S) matrix have?

Accepted Answer

Introduction: S-parameters describe how incident and reflected waves relate at each port of an RF network. A port is matched when a wave incident on that port produces no reflection. This condition is compactly expressed in the S-matrix as a requirement on its diagonal entries. Given Data / Assumptions: Linear, time-invariant microwave network with N ports. Reference impedances chosen equal to the intended match (typically 50 Ω). Definition: matched port → Sii = 0. Concept / Approach: The i-th diagonal entry Sii equals the reflection coefficient seen looking into port i when all other ports are terminated in their reference impedances. “Matched at all ports” means S11 = S22 = … = SNN = 0. Off-diagonal entries determine transmission/isolation properties, but they do not affect the matched condition itself. Step-by-Step Solution: 1) Matched at port i ⇒ Γi = 0 under standard terminations.2) In S-parameter notation, Γi = Sii.3) Therefore, all ports matched ⇒ all Sii = 0. Verification / Alternative check: For a Wilkinson power divider designed for 50 Ω at all ports, the measured |S11|, |S22|, |S33| are near zero across the design band, illustrating the condition. Why Other Options Are Wrong: Equal non-zero diagonals still imply reflections; “purely complex” is irrelevant; Hermitian pertains to reciprocity and losslessness only under special cases; unitarity (lossless) does not imply zero diagonals. Common Pitfalls: Confusing “lossless” (unitary S) with “matched.” A device can be lossless yet unmatched, or matched yet lossy. Final Answer: all diagonal elements are zero.

Question 18

Earth’s Magnetic Field vs. Radio Wave Polarization  In ionospheric propagation and magneto-ionic theory, the influence of Earth’s magnetic field on a passing radio wave depends on their relative orientation. The effect is minimum when which of the f…

Accepted Answer

Introduction / Context: As a radio wave travels through the ionosphere, the presence of Earth’s magnetic field causes magneto-ionic effects such as Faraday rotation and mode splitting into ordinary and extraordinary waves. The strength of this interaction depends on how the wave’s polarization is oriented relative to Earth’s magnetic flux lines. This question asks for the orientation that minimizes the magnetic-field influence on the wave. Given Data / Assumptions: Plane radio wave passing through a magnetized plasma (ionosphere). Comparison is about relative orientation between the wave’s electric field vector and Earth’s magnetic field vector. Frequency is high enough for ionospheric propagation relevance (HF/VHF/UHF as applicable). Concept / Approach: In magneto-ionic theory, the coupling that leads to polarization rotation is strongest when the field components align to drive electron motion along or against the ambient magnetic field. When the wave’s electric field is perpendicular to Earth’s magnetic field, the Lorentz-force-related coupling for motion along the field is minimized, reducing the net magneto-ionic effect (e.g., minimized Faraday rotation rate for that polarization orientation). Step-by-Step Solution: 1) Identify vectors: E_wave (electric field) and B_Earth (geomagnetic field).2) Magneto-ionic interaction depends on the projection of E-induced motion along B_Earth.3) If E_wave ⟂ B_Earth, projection is minimized, so magnetic-field-induced effects are smallest.4) Therefore, the minimum effect occurs when the wave’s electric field is at right angles to Earth’s magnetic flux lines. Verification / Alternative check: Derivations of Faraday rotation rate include a factor that depends on the angle between the propagation/polarization and the geomagnetic field; the rate decreases as the relevant component reduces, reaching minimum when the polarization is orthogonal to B_Earth for a given path. Why Other Options Are Wrong: Same direction (A) or opposite (C) maximize the component along B and therefore increase magneto-ionic effects; (D) cannot be correct since (C) is not minimizing; (E) references the wave’s magnetic field, not its electric field, and does not directly state the minimizing condition asked. Common Pitfalls: Confusing the wave’s magnetic field with the electric field; ignoring that polarization is defined by the electric field vector for most practical antenna considerations. Final Answer: the electric field lines of radio waves and flux lines of Earth’s magnetic field are at right angles

Question 19

Bolometric Microwave Power Measurement  In a microwave power meter using a bolometer, which physical quantity is intentionally varied with absorbed RF power to indicate power level?

Accepted Answer

Introduction / Context: Bolometric power meters are classic instruments for measuring microwave power with high accuracy. They convert RF power into heat, then sense a resulting change in an electrical property to infer the incident power. Given Data / Assumptions: Bolometer element (thermistor or barretter) is used as the sensing element. Absorbed RF power raises the temperature of the element. A bridge or bias circuit monitors the change to produce a calibrated reading. Concept / Approach: The defining characteristic of a bolometer is that its electrical resistance varies predictably with temperature. When RF power is absorbed, temperature increases and the resistance changes. By keeping the bridge in balance (or reading the resistance shift), one can determine the absorbed power. Therefore, the measured quantity that varies with power is resistance, not inductance or capacitance. Step-by-Step Solution: 1) RF power → heat in the sensing element.2) Temperature rise → resistance change (R(T)).3) Bridge or feedback loop converts resistance change to an indication of input power. Verification / Alternative check: Thermistor bolometers have negative temperature coefficient; barretter types have positive temperature coefficient. Both rely on resistance-temperature dependence. Why Other Options Are Wrong: Inductance/capacitance (A,C) are not the primary sensing mechanisms in bolometers; cavity dimension (D) belongs to resonant-cavity power methods; Q-factor (E) also refers to resonator-based measurements, not bolometric sensing. Common Pitfalls: Confusing bolometric methods with thermocouple or diode detectors; assuming frequency-selective cavities when the device is broadband absorptive. Final Answer: resistance with absorption of power

Question 20

Primary (Per-Unit-Length) Constants of a Transmission Line  Which set correctly lists the four primary constants that define a uniform transmission line?

Accepted Answer

Introduction / Context: Uniform transmission lines are characterized by parameters per unit length that determine wave propagation, attenuation, and impedance. These constants appear in the telegrapher’s equations and govern behavior across frequency. Given Data / Assumptions: Uniform, linear transmission line model. Low-loss and high-loss cases are both encompassed by the same set of constants. We are looking for a complete set of primary constants. Concept / Approach: The four primary constants are: R (series resistance per unit length), L (series inductance per unit length), G (shunt conductance per unit length), and C (shunt capacitance per unit length). Together they determine the propagation constant γ and characteristic impedance Z0 through standard relations. Step-by-Step Solution: 1) Identify series parameters: R and L model conductor loss and magnetic energy storage.2) Identify shunt parameters: G and C model dielectric loss and electric energy storage.3) Conclude the complete set is {R, L, G, C} per unit length. Verification / Alternative check: γ = sqrt((R + jωL) * (G + jωC)) and Z0 = sqrt((R + jωL) / (G + jωC)) both require all four quantities to be generally valid. Why Other Options Are Wrong: Pairs or triplets (A, B, C) omit necessary terms; Z0 (E) is derived from the primary constants and is not itself a primary constant. Common Pitfalls: Assuming “low-loss” means G = 0; while often small, it is part of the general model. Final Answer: R, L, G and C

Microwave Communication Questions