Question 1

Flip-flop timing integrity: What is a race condition in the context of an edge-triggered flip-flop and its input timing?

Accepted Answer

Introduction / Context: A race condition in sequential logic occurs when signals that should be stable during the sampling event change too close to the active clock edge, potentially violating setup or hold requirements and causing unpredictable outputs or metastability. Given Data / Assumptions: Edge-triggered flip-flop with defined setup time ts and hold time th. Inputs intended to be stable around the sampling edge. Clock with finite transition time. Concept / Approach: When inputs transition during the aperture window around the clock edge, internal latches can capture inconsistent states, leading to raced or metastable outcomes. Ensuring ts and th margins avoids races. Step-by-Step Solution: 1) Identify the aperture: from (edge − ts) to (edge + th).2) If input changes inside this window, the device may not resolve deterministically.3) Design practice: constrain combinational delays and synchronize asynchronous signals.4) Add proper timing constraints and CDC synchronizers where needed. Verification / Alternative check: Static timing analysis should show positive slack for ts and th; oscilloscope capture reveals glitches if violated. Why Other Options Are Wrong: Changing slightly before the edge: If it is earlier than ts, it is acceptable. Changing slightly after the edge: If it is later than th, it is acceptable. Static and unchanged: This is the desired condition, not a race. Common Pitfalls: Feeding asynchronous inputs directly to synchronous logic without synchronization, and neglecting clock skew while budgeting ts/th. Final Answer: The inputs to a trigger device are changing at the same time as the active trigger edge.

Question 2

Flip-flop timing parameters: Which parameters specify the propagation delay from the clock (Cp) input to the Q output?

Accepted Answer

Introduction / Context: Datasheets list several timing parameters for sequential devices. Correctly identifying propagation delays from clock to output is essential for timing closure and performance prediction. Given Data / Assumptions: Edge-triggered flip-flops with defined delays. tPHL and tPLH denote propagation from input to output with polarity. Other parameters include setup/hold, pulse width, and maximum frequency. Concept / Approach: Propagation delay measures time for an output transition after a triggering input event. For clocked devices, the relevant labels are tPLH (LOW-to-HIGH at output) and tPHL (HIGH-to-LOW at output) after a defined clock edge. Step-by-Step Solution: 1) Identify the cause: active clock edge at Cp.2) Map to effects: Q transitions occur after a delay.3) Read tPLH/tPHL as the delays for the two transition polarities. Verification / Alternative check: Scope measurements of Q relative to Cp confirm datasheet values within tolerances. Why Other Options Are Wrong: ts, th: Setup and hold requirements, not propagation delay. tw(L), tw(H): Minimum pulse widths, not output delays. fmax: Maximum toggling frequency, a derived bound, not a delay parameter. Common Pitfalls: Confusing tpd with setup/hold and ignoring process/voltage/temperature effects on delays. Final Answer: tPHL, tPLH

Question 3

Pull-down resistor caution: What is the primary concern when biasing an input with a pull-down resistor?

Accepted Answer

Introduction / Context: Pull-down resistors bias an input to logic LOW when undriven. While useful, they can increase steady-state current draw and power loss, especially when the node is driven HIGH for significant time. Given Data / Assumptions: VCC is present and the signal may be HIGH for long durations. Pull-down connects the node to ground through a resistor. Input leakage is small but nonzero. Concept / Approach: When the node is driven HIGH, current flows from VCC through the driver into the resistor to ground, dissipating power P = V^2 / R approximately (neglecting driver drop). Choosing too small a resistor leads to unnecessary power and thermal stress. Step-by-Step Solution: 1) Consider the worst case: node HIGH continuously with pull-down to ground.2) Compute power in resistor: P ≈ (VCC^2) / R.3) Select R large enough to limit current yet small enough to provide noise immunity when undriven.4) Evaluate logic-family input currents to set the allowable R range. Verification / Alternative check: Thermal check of resistor and supply current measurements validate the choice. Why Other Options Are Wrong: Low power dissipation: The concern is the opposite—excess dissipation. Keeps a floating terminal LOW: That is a benefit, not a concern. False triggering: Possible with poor values, but the primary design worry is wasted power. Common Pitfalls: Copying pull-up values without considering asymmetry, using very small resistances, and ignoring battery life impacts. Final Answer: the high power dissipation of the resistor

Question 4

J-K versus S-R flip-flop: What is the key advantage of the J-K flip-flop when compared to the S-R latch/flip-flop?

Accepted Answer

Introduction / Context: The S-R latch has an invalid input condition when both S and R are HIGH simultaneously. The J-K flip-flop extends this behavior to allow a defined toggle action when both inputs are asserted, making it more versatile. Given Data / Assumptions: J-K is edge-triggered in most logic families. Inputs J = 1 and K = 1 cause toggling on each active clock edge. S-R with S = R = 1 is invalid or forbidden. Concept / Approach: The toggle state enables divide-by-2 counters and flexible state machines without additional gating to avoid illegal states, simplifying designs. Step-by-Step Solution: 1) Map inputs: J=1, K=0 → set; J=0, K=1 → reset; J=K=0 → hold; J=K=1 → toggle.2) Compare to S-R: S=R=1 is undefined; hardware must prevent it.3) Conclude advantage: J-K provides a clean, useful toggle mode. Verification / Alternative check: Construct a divide-by-2 using a single J-K with J=K=1; observe Q toggling each clock. Why Other Options Are Wrong: Much faster: Speed depends on implementation, not a fundamental advantage. No propagation delay problems: All real devices have propagation delays. Two outputs: Many flip-flops provide Q and not-Q; this is not unique to J-K. Common Pitfalls: Assuming J-K eliminates all illegal transitions automatically; race-around must be mitigated by edge triggering or master-slave design. Final Answer: The J-K flip-flop has a toggle state.

Question 5

Digital electronics — input jitter and false switching mitigation In digital systems, which circuit specifically overcomes erratic switching caused by noisy or jittery input edges by introducing hysteresis?

Accepted Answer

Introduction / Context: Schmitt trigger circuits are widely used in digital electronics to clean up slow, noisy, or jittery input signals. They provide two distinct threshold voltages (one for rising and another for falling inputs), a behavior known as hysteresis. This prevents multiple unwanted transitions when an input hovers near a single threshold, thereby ensuring stable logic-level decisions. Given Data / Assumptions: We are dealing with a digital input that exhibits jitter or noise around its switching point. Goal is to avoid false triggering or multiple transitions near the threshold. Standard logic families expect clean, fast transitions. Concept / Approach: The key concept is hysteresis. A Schmitt trigger implements two thresholds: V_T+ for LOW to HIGH transitions and V_T- for HIGH to LOW transitions. Once the input crosses V_T+, the output switches and will not switch back until the input falls below V_T-. This input hysteresis window filters slow edges and noise. Step-by-Step Solution: 1) Identify the problem: input jitter near a single threshold causes repeated toggling.2) Require a device with separate rising and falling thresholds to add immunity.3) Recognize that a Schmitt trigger provides hysteresis by design.4) Conclude the correct choice for robust switching is the Schmitt trigger. Verification / Alternative check: Oscilloscope measurements show that without hysteresis, a noisy slow edge produces multiple transitions. With a Schmitt trigger, the output flips once on crossing V_T+ and remains stable until the signal clearly crosses V_T-, eliminating chatter. Why Other Options Are Wrong: Astable multivibrator: generates a free-running square wave, not a noise-immune input buffer. Monostable multivibrator: produces one pulse per trigger, not continuous edge cleanup. Bistable multivibrator: a latch or flip-flop with no inherent hysteresis window on an analog input. Common Pitfalls: Confusing edge shaping (debouncing) with pulse generation. Monostables can time-limit responses but do not fix a noisy threshold. The Schmitt trigger specifically addresses noisy transitions via hysteresis. Final Answer: Schmitt trigger

Question 6

TTL NAND driving an LED indicator LOW (active-LOW sink configuration) A standard TTL NAND output sinks current through a 470 Ω series resistor and LED to VCC = 5 V. What current flows when the LED is ON (output LOW)?

Accepted Answer

Introduction / Context: When a TTL gate output pulls LOW, it can sink current from a load tied to VCC. For an LED indicator with a series resistor connected to VCC, the LED current is set by the voltage across the resistor, which depends on VCC, the LED forward drop, and the LOW-level output voltage of the gate (VOL). The question tests understanding of active-LOW LED driving and basic current calculation. Given Data / Assumptions: Logic family: standard TTL NAND, VCC ≈ 5 V. Series resistor: R = 470 Ω in series with the LED, tied to VCC. Gate output is LOW when LED is ON, so current flows from VCC → resistor → LED → gate output to ground. Use typical textbook values for LED forward drop and VOL to obtain the closest numerical option. Concept / Approach: Ohm's law sets current I = V_R / R, where V_R is the voltage across the resistor. V_R = VCC − V_LED − VOL. Typical small red LED forward drop is about 2.0 V to 2.2 V under a few milliamps. A TTL VOL under modest sink currents is typically a few hundred millivolts. The closest standard value among the options generally corresponds to around 5 mA to 6 mA. Step-by-Step Solution: 1) Assume VCC = 5.0 V.2) Assume V_LED ≈ 2.2 V (typical) and VOL ≈ 0.3 V (representative for TTL at a few mA).3) Compute resistor voltage: V_R = 5.0 − 2.2 − 0.3 = 2.5 V.4) Compute current: I = V_R / R = 2.5 / 470 ≈ 0.00532 A = 5.32 mA. Verification / Alternative check: Trying nearby typical values (for example, V_LED between 2.0 and 2.3 V and VOL between 0.2 and 0.4 V) yields currents in the 5 mA to 6.5 mA range. The given options include 5.32 mA, which matches the calculation using mainstream assumptions. Why Other Options Are Wrong: 7.02 mA: implies a much lower LED drop or VOL, not typical at this operating point. 8.51 mA: requires an unrealistically low combined drop across LED and VOL for 470 Ω. 10.63 mA: far too high for the stated component values and typical TTL levels. Common Pitfalls: Forgetting to include VOL. Assuming the LED is tied to ground rather than VCC in an active-LOW sink configuration leads to the wrong reference polarities. Final Answer: 5.32 mA

Question 7

Use-case of a Schmitt trigger in practical digital circuits Which application best illustrates the advantage of hysteresis for cleaning noisy, bouncing inputs?

Accepted Answer

Introduction / Context: Mechanical pushbuttons and switches do not make a clean single transition; they bounce, producing rapid ON-OFF edges. A Schmitt trigger adds hysteresis, ensuring only one clean digital transition despite noisy or bouncing inputs. This question checks recognition of the most common real-world use-case for Schmitt triggers. Given Data / Assumptions: Input is from a mechanical switch that exhibits contact bounce. We want a single, stable digital-level transition at the logic output. Hysteresis eliminates multiple triggers near the threshold. Concept / Approach: Schmitt triggers implement two thresholds: V_T+ and V_T-. An input must cross V_T+ to be registered as HIGH and later fall below V_T- to register LOW. This window rejects intermediate chatter due to bounce or small noise spikes. Step-by-Step Solution: 1) Identify the symptom: multiple spurious transitions from a bouncing switch.2) Choose an input conditioning method that requires a clear overdrive past a threshold window.3) A Schmitt trigger naturally provides hysteresis to clean the signal.4) Therefore, the best application is switch debouncing. Verification / Alternative check: Compare scope traces: a raw switch shows multiple edges; after a Schmitt trigger, only one clean edge remains per actuation. Why Other Options Are Wrong: Racer: not a defined digital building block and does not address bounce. Astable oscillator: generates periodic signals rather than debouncing an input. Transition pulse generator: shapes edges but does not inherently solve bounce without hysteresis. Common Pitfalls: Confusing simple RC filtering with guaranteed single-edge switching. RC alone may still allow chatter; hysteresis provides deterministic thresholds. Final Answer: switch debouncer

Question 8

Flip-flop behavior near the active clock edge What is the condition called when flip-flop inputs change at the same instant the active clock edge occurs, risking an indeterminate or unintended output?

Accepted Answer

Introduction / Context: Flip-flops sample inputs at an active clock edge. If inputs change too close to that edge, setup or hold time requirements can be violated, leading to uncertain behavior. Many curricula refer to the resulting hazard as a race or race condition, closely tied to metastability in practical devices. Given Data / Assumptions: Inputs are changing as the clock edge arrives. Device requires finite setup and hold time margins. Violation of these margins triggers unpredictable or oscillatory outcomes. Concept / Approach: If data transitions occur within the aperture around the sampling edge, the storage nodes can enter a metastable state or produce unexpected toggling. This conflict between changing data and the latching action is commonly described as a race condition in many introductory texts on sequential logic. Step-by-Step Solution: 1) Recognize that flip-flops need stable inputs around the edge (setup and hold).2) Inputs changing at the sampling instant violate those requirements.3) Such timing leads to a race between data settling and latching.4) Hence, the condition is termed racing (race condition). Verification / Alternative check: Manufacturers specify setup and hold in datasheets. Timing analysis ensures input edges avoid the sampling aperture; otherwise, failures consistent with racing or metastability appear in hardware tests. Why Other Options Are Wrong: Toggling: normal behavior of a T or JK flip-flop, not a timing hazard description. Slave loading: not the standard name for this hazard. Pulse timing: generic phrase, not the accepted term for the race issue. Common Pitfalls: Using asynchronous input signals without synchronization, or ignoring clock-to-output, setup, and hold in timing closure. Final Answer: racing

Question 9

Power-up reset generation in digital design: A simple series R–C network can be used to generate an automatic power-up reset pulse that initializes digital logic when supply voltage rises. Evaluate this statement for typical TTL/CMOS systems.

Accepted Answer

Introduction / Context: At power-on, digital systems must come up in a known state. A common, low-cost approach is to derive a brief reset pulse from the power ramp using a resistor–capacitor (R–C) network and a buffer or Schmitt trigger. This question checks practical understanding of power-up reset techniques. Given Data / Assumptions: Supply voltage rises from 0 V to its nominal value. An R–C network is connected so that the capacitor initially holds a logic level that asserts reset. As the capacitor charges through the resistor, the reset signal de-asserts after a delay. Downstream logic accepts an asynchronous reset pulse of sufficient width and level. Concept / Approach: An R–C differentiates the supply ramp to create a time-limited signal. With proper polarity and thresholding (often via a Schmitt trigger input), the node asserts reset during the early portion of the ramp and then releases reset as the capacitor charges, ensuring a defined initialization interval. The reset width τ is roughly R*C (subject to logic thresholds and VCC ramp rate). Step-by-Step Solution: Choose R and C so that τ = R*C comfortably exceeds the minimum reset time of the logic.Connect so that at power-up the capacitor voltage starts near 0 V, producing an active reset.As VCC rises and the capacitor charges, the node crosses the release threshold, de-asserting reset.Optionally pass through a Schmitt trigger for clean edges and immunity to slow ramps. Verification / Alternative check: Many microcontroller reference designs show R–C reset or recommend supervisor ICs that formalize this behavior with precise thresholds and timing. Why Other Options Are Wrong: “Incorrect” ignores long-standing practice. The AC-only and 15 V restrictions are irrelevant; R–C resets are common at 3.3 V or 5 V with proper component values. Common Pitfalls: Using slow ramps without hysteresis; choosing τ too short; failing to ensure reset meets minimum pulse width across temperature and tolerance. Final Answer: Correct

Question 10

Schmitt trigger characteristics: A Schmitt trigger employs positive feedback to create two distinct switching thresholds (hysteresis), thereby cleaning up noisy or slowly changing input signals. Judge the accuracy of this statement.

Accepted Answer

Introduction / Context: Schmitt triggers are ubiquitous in digital interfaces where inputs may rise or fall slowly, or where noise could cause spurious toggling. By adding hysteresis, these devices enforce clean transitions and reduce chatter at threshold. Given Data / Assumptions: Input may be noisy or have a slow slew rate. Schmitt trigger topology includes positive feedback. Two thresholds exist: one for rising input, one for falling input. Concept / Approach: Positive feedback shifts the effective switching threshold depending on the current output state. This produces hysteresis: V_T+ (rising) and V_T− (falling), with V_T+ > V_T−. The gap between these thresholds filters input noise within that band and ensures the output changes state only once per valid crossing. Step-by-Step Solution: Model the trigger as a comparator/inverter with resistive feedback.Derive that the input threshold depends on output level through feedback division.Observe distinct rising vs falling trip points.Conclude that hysteresis stems from the intentional positive feedback. Verification / Alternative check: Datasheets for logic families with Schmitt inputs (e.g., 74HC14) specify separate V_T+ and V_T−, confirming hysteresis behavior. Why Other Options Are Wrong: “Incorrect” disregards the defining feature. The behavior holds for both inverting and noninverting forms and across low-voltage logic, not only above arbitrary voltages. Common Pitfalls: Assuming hysteresis means memory beyond thresholds; in fact, it is immediate state-dependent thresholding. Overlooking that hysteresis width affects noise immunity and timing. Final Answer: Correct

Question 11

TTL supply requirement — fact check: Transistor–Transistor Logic (TTL) devices require a regulated supply near 5 V (nominal), not 8.0 V. Evaluate the statement: “TTL requires a constant supply voltage of 8.0 V.”

Accepted Answer

Introduction / Context: Correct supply voltage knowledge is essential to avoid damaging logic devices. TTL families (74xx, 74LS, 74ALS, etc.) have long standardized on a 5 V supply, with defined noise margins and logic thresholds around that nominal value. Given Data / Assumptions: Standard TTL operates at VCC ≈ 5 V (typical 4.75–5.25 V tolerance for many families). Absolute maximum ratings prohibit substantially higher continuous supply voltages. Some CMOS families (e.g., 4000B) tolerate wider ranges, but that is not TTL. Concept / Approach: The proposed 8.0 V requirement is incompatible with TTL design. Transistor biasing, input thresholds, and output levels are engineered for 5 V rails. Applying 8 V risks exceeding maximum ratings and destroying the device. Specialized translators or regulator stages are used when interfacing to other voltages. Step-by-Step Solution: Identify family: TTL → nominal 5 V.Check datasheet ranges: typically around 4.75–5.25 V for standard parts.Compare to claim: 8.0 V is far outside operating limits.Therefore the statement is false. Verification / Alternative check: Datasheets for 74LS00, 74ALS00, etc., specify VCC = 5 V nominal. None require 8 V. Why Other Options Are Wrong: Open-collector outputs do not change core supply needs. “Military-grade” or temperature ratings do not redefine VCC to 8 V. Common Pitfalls: Confusing TTL with certain analog or CMOS families; misreading absolute maxima as operating points. Final Answer: Incorrect

Question 12

Pull-up and pull-down resistors — purpose check: Engineers use pull-ups to hold a floating input HIGH and pull-downs to hold a floating input LOW. Assess the statement: “Pull-up resistors and pull-down resistors are used to keep a floating terminal …

Accepted Answer

Introduction / Context: Reliable logic requires defined input levels. Floating inputs can drift, oscillate, or pick up noise. Pull-up and pull-down resistors provide a default logic level when no active driver is present, preventing undefined behavior and excess power consumption. Given Data / Assumptions: “Floating” means high-impedance, no strong driver. Pull-up connects the node to VCC through a resistor. Pull-down connects the node to GND through a resistor. Active drivers can override the weak pull. Concept / Approach: A pull-up biases a node HIGH; a pull-down biases it LOW. The statement claims both are used to keep a floating terminal HIGH, which is incorrect since pull-downs explicitly bias LOW. Correct usage depends on the logic polarity and desired idle state (e.g., active-low inputs typically use pull-ups). Step-by-Step Solution: Identify desired default: HIGH → use pull-up; LOW → use pull-down.Confirm the statement lumps both as keeping HIGH → incorrect.Note special cases (open-drain/open-collector) still use pull-ups for HIGH; pull-downs are not for HIGH.Therefore, the claim is false. Verification / Alternative check: MCU application notes illustrate input mode with pull-ups for buttons to ground (idle HIGH) and pull-downs for buttons to VCC (idle LOW). Why Other Options Are Wrong: “Correct” contradicts definitions. Open-drain outputs indeed require pull-ups for HIGH, but that does not make pull-downs a means to hold HIGH. Schmitt triggers improve noise margins but do not change pull-down polarity. Common Pitfalls: Leaving CMOS inputs floating; choosing pull values too strong (wasting power) or too weak (slow edges, noise-susceptible). Final Answer: Incorrect

Question 13

Phototransistor operation — control mechanism: A phototransistor’s conduction, and thus its output state in interfacing circuits, is governed by the presence or absence (and intensity) of incident light acting as an effective base drive. Evaluate th…

Accepted Answer

Introduction / Context: Opto-sensors such as phototransistors are common in digital input conditioning, opto-isolators, and light detection. Understanding how light influences their conduction clarifies why pull-ups, biasing, and thresholding are arranged as they are in interface circuits. Given Data / Assumptions: A phototransistor responds to photons that generate carriers, effectively providing base current. Many phototransistors omit a discrete base lead; light acts as the stimulus. External biasing and load resistors translate collector current into a voltage. Concept / Approach: Incident light increases the effective base drive, raising collector current and changing the voltage across the load. In common-emitter configurations with a pull-up, more light typically pulls the output node LOW (more current through the transistor), while in other topologies the logic polarity can be inverted. The essence remains: light presence/absence controls conduction. Step-by-Step Solution: Model the device as a BJT with optically induced base current.More light → more carriers → higher collector current.Translate current to voltage with the load to obtain a logic-level signal.Set thresholds so ambient variations do not cause false triggers. Verification / Alternative check: Datasheets specify “collector current vs irradiance” curves confirming monotonic response to light intensity. Why Other Options Are Wrong: Requiring an external base lead is unnecessary; many packages are base-less. Photodiodes differ in operation (reverse-biased junction current) though both are light-sensitive; the statement correctly concerns phototransistors. Common Pitfalls: Ignoring ambient light; failing to add hysteresis; misinterpreting logic polarity based on wiring configuration. Final Answer: Correct

Question 14

Terminology accuracy — “race condition” vs setup time: The statement defines “race condition” as the time a valid signal must be present at a flip-flop input before the active clock edge (i.e., setup time). Decide whether this definition is correct …

Accepted Answer

Introduction / Context: Precise terminology matters for debugging timing problems. Setup time is the required interval that input data must be stable before a clock edge to guarantee correct sampling. A race condition is a different concept, typically referring to unpredictable behavior when signal paths compete or evaluation order affects outcome. Given Data / Assumptions: Flip-flop timing parameters include setup time (t_setup) and hold time (t_hold). Race conditions arise in asynchronous logic, level-sensitive latches, or poorly constrained paths where signals “race” to influence state. The statement equates “race condition” with setup time definition. Concept / Approach: Setup time is a deterministic device parameter: data must be valid for at least t_setup before the clock. A race condition, in contrast, is system-level: two or more transitions compete such that the final state depends on the relative, often uncontrollable, arrival times. Examples include “race-around” in level-sensitive JK latches or hazard-induced glitches propagating through asynchronous networks. Step-by-Step Solution: Identify the statement’s definition → it describes setup time.Contrast with true race condition → outcome depends on path delays and ordering, not a fixed specification.Conclude: the statement mislabels setup time as race condition.Therefore, it is incorrect usage of terminology. Verification / Alternative check: Device datasheets define t_setup/t_hold; digital design texts discuss race conditions in asynchronous circuits and latch transparency, not as a fixed pre-clock interval. Why Other Options Are Wrong: Limiting correctness to latches or FIFO pointers does not fix the definitional error; those domains can have races, but races are not synonymous with setup time. Common Pitfalls: Confusing metastability (violating setup/hold) with races; using “race” to describe any timing failure instead of the specific phenomenon of competing paths. Final Answer: Incorrect

Question 15

Schmitt trigger fundamentals — A transfer function (input–output) graph for a Schmitt trigger illustrates its most important specifications, including thresholds and hysteresis width. Evaluate the statement.

Accepted Answer

Introduction / Context: A Schmitt trigger is a comparator with positive feedback that exhibits hysteresis. Its most important practical specifications are the two switching thresholds (upper and lower), the hysteresis width, and the relationship between input voltage and output state. A transfer function graph (sometimes called a voltage transfer characteristic) captures these facts in a single, intuitive plot. Given Data / Assumptions: The device is a Schmitt trigger (inverting or non-inverting topology). Transfer function graph plots V_out versus V_in and indicates threshold points. Supply rails and logic output levels are within normal operating conditions. Concept / Approach: The Schmitt trigger’s hallmark is hysteresis: the rising-input threshold V_T+ and the falling-input threshold V_T− are different values, separated by a hysteresis width H = V_T+ − V_T−. A transfer graph shows the direction-dependent switching: as V_in rises past V_T+, the output snaps to the opposite rail; as V_in falls past V_T−, it snaps back. This visual directly communicates noise immunity and the input margin that prevents chatter near a single threshold. Step-by-Step Solution: Identify V_T+ on the rising trajectory where the output toggles.Identify V_T− on the falling trajectory where the output toggles back.Compute hysteresis width H = V_T+ − V_T−; this is the noise band immune to spurious switching.Relate output levels to supply rails (e.g., logic HIGH and LOW). Verification / Alternative check: Datasheets list V_T+, V_T− and often include a transfer plot; the plot matches tabulated thresholds and shows the “loop” that embodies hysteresis. Why Other Options Are Wrong: Incorrect: Ignores that thresholds and hysteresis are literally read off the transfer curve.Only true for non-inverting types: Inverting and non-inverting both have transfer curves that show thresholds.True only at 25 °C: While values vary with temperature, the graph form and what it illustrates remain valid. Common Pitfalls: Confusing a simple comparator (single threshold) with a Schmitt trigger (two thresholds).Reading a DC transfer plot as if it were a dynamic timing diagram; it is quasi-static behavior. Final Answer: Correct

Question 16

Switch bounce in digital systems — Evaluate the claim: “There is no way to eliminate the effects of switch bounce.” Provide the best engineering perspective.

Accepted Answer

Introduction / Context: Mechanical switches do not close once and stay closed; their contacts physically bounce, producing multiple rapid transitions within a few milliseconds. If these transitions are not handled, digital circuits may interpret several presses or releases instead of one. The question asks whether bounce effects can be eliminated from the perspective of a system designer. Given Data / Assumptions: Bounce time typically spans 1–20 milliseconds depending on switch type. Digital inputs have finite threshold and may trigger on spurious edges. Designers can add hardware or firmware conditioning stages. Concept / Approach: While you cannot stop the physics of contacts bouncing, you can eliminate its effects on the digital domain using debouncing. Hardware methods include RC low-pass filters, Schmitt trigger buffers, dedicated debouncer ICs, or synchronous sampling with a flip-flop chain. Software methods sample at intervals and accept a change only if it remains stable for a set debounce interval (e.g., 10–30 ms). The result is a single clean transition recognized by the logic, effectively eliminating bounce effects at the system level. Step-by-Step Solution: Apply RC + Schmitt trigger to convert bounces into a monotonic edge.Alternatively, sample the switch in firmware; require stability for N consecutive samples before state change.Synchronize the signal into the system clock domain to avoid metastability.Verify with a scope or logic analyzer that only one clean edge reaches the downstream logic. Verification / Alternative check: Experimentally, adding a 10 kΩ/100 nF network and a Schmitt buffer reliably yields a single transition for most pushbuttons. Why Other Options Are Wrong: Correct: Suggests inevitability; ignores standard hardware/software debouncing solutions.Only true for toggle switches: All mechanical switches bounce; mitigation applies broadly.True unless Schmitt triggers are used: Schmitt triggers help but often are paired with RC or firmware logic for best results. Common Pitfalls: Feeding raw switch signals straight into edge-triggered inputs (counters, clocks).Insufficient debounce interval causing occasional double-detects. Final Answer: Incorrect

Question 17

Linear regulator thermal insight — Evaluate: “A 7805 will get very hot if your circuit draws more than 0.5 A.” Consider power dissipation and input–output voltage.

Accepted Answer

Introduction / Context: The 7805 is a linear regulator that outputs approximately 5 V. Its temperature rise depends on power it must dissipate as heat, not on current alone. The statement claims any load above 0.5 A guarantees excessive heating, which oversimplifies how linear regulators work. Given Data / Assumptions: Output voltage V_out ≈ 5 V. Input voltage V_in can vary (e.g., 6–12 V or more). Power dissipation P = (V_in − V_out) * I_load. Thermal behavior depends on package, heatsink, airflow, and ambient temperature. Concept / Approach: A linear regulator drops the excess voltage across itself. The larger the drop and the higher the load current, the higher the dissipation. Therefore, whether it “gets very hot” at 0.5 A depends on V_in and thermal design. For example, with V_in = 5.5 V and I = 0.5 A, P = (5.5 − 5) * 0.5 = 0.25 W, often manageable. But with V_in = 12 V and I = 0.5 A, P = 3.5 W, which will cause rapid heating without a substantial heatsink. Step-by-Step Solution: Compute dissipation: P = (V_in − 5) * I.Estimate junction rise: T_rise ≈ P * θ_JA (package thermal resistance).Decide on heatsinking: If T_j exceeds safe limits, add heatsink or reduce (V_in − V_out).Consider alternatives: A buck regulator for large voltage drops and high current. Verification / Alternative check: Measure case temperature at various V_in; results show strong dependence on voltage drop, not current alone. Why Other Options Are Wrong: Correct: Not universally true; ignores role of V_in and heatsinking.Correct only when Vin ≥ 9 V: Heating can be acceptable even above 9 V with light current; or problematic below 9 V with heavy current.Correct for TO-92 packages only: 7805s in TO-220 dissipate more, but heating still depends on P, not only package. Common Pitfalls: Using a high V_in from a wall adapter and expecting low heat at large currents.Ignoring dropout voltage and thermal shutdown behavior. Final Answer: Incorrect

Question 18

Flip-flop timing requirements — Evaluate: “The input levels to a flip-flop must be maintained for a minimum time period both before and after the active clock edge.”

Accepted Answer

Introduction / Context: Edge-triggered flip-flops capture input data at a particular clock edge. To ensure reliable operation, the data and control inputs must be stable around that edge. These time windows are known as setup time (before the edge) and hold time (after the edge). Given Data / Assumptions: We consider synchronous data inputs (e.g., D for D-FF, J/K under synchronous use). “Active edge” means rising or falling edge that triggers capture. Device timing parameters come from datasheets and vary by technology. Concept / Approach: Setup time t_setup: the signal must be stable for at least t_setup before the clock edge. Hold time t_hold: the signal must remain stable for at least t_hold after the edge. Violating either can cause metastability or incorrect data capture. Therefore, the statement accurately describes a universal requirement for synchronous inputs to flip-flops. Step-by-Step Solution: Ensure data meets t_setup relative to the clock edge.Ensure data meets t_hold following the clock edge.Budget clock skew and propagation delays to maintain margins across the design.Verify in timing analysis tools and with oscilloscope measurements where applicable. Verification / Alternative check: Static timing analysis explicitly checks setup/hold constraints between registers; silicon failures often trace to violations. Why Other Options Are Wrong: Incorrect: Would ignore well-established timing constraints.Applies only to asynchronous inputs: Asynchronous inputs use different constraints (like pulse width, recovery/removal), not setup/hold.True only for D flip-flops: All clocked flip-flops impose setup/hold on their synchronous data paths. Common Pitfalls: Forgetting clock skew/jitter reduces effective timing margins.Confusing asynchronous set/reset with synchronous data timing. Final Answer: Correct

Digital Design Questions