Question 1

Ideal inductor power concept: For a pure (ideal) inductor driven by a sinusoidal source, how does average power dissipation relate to operating frequency?

Accepted Answer

Introduction / Context: This question distinguishes reactive energy storage from real power dissipation. In AC circuits, inductors and capacitors exchange energy with the source but do not dissipate average power if they are ideal (no resistance). Understanding this helps in power factor correction and reactive component selection. Given Data / Assumptions: Inductor is ideal: no series resistance, no core losses. Source is sinusoidal steady state. We ask about average (real) power, not instantaneous energy exchange. Concept / Approach: For a pure inductor, voltage and current are 90 degrees out of phase. Instantaneous power p(t) = v(t) * i(t) is alternating positive and negative; the average over a full cycle is zero. Frequency changes the reactive impedance X_L = 2 * pi * f * L, affecting current magnitude for a given voltage, but average real power in the ideal case remains zero regardless of frequency. Step-by-Step Solution: Use X_L = 2 * pi * f * L to see current scaling with frequency.Recognize phase shift: current lags voltage by 90 degrees in an ideal inductor.Average real power = 0 for ideal reactive elements; only reactive power oscillates.Hence, no “increasing” or “decreasing” trend with frequency for average dissipation. Verification / Alternative check: Include small series resistance R_s for realism: P_avg ≈ I^2 * R_s. Here, frequency affects I via X_L, but the strictly ideal component has R_s = 0, giving P_avg = 0 at any f. Why Other Options Are Wrong: (a) and (c) claim a monotonic trend; not true for an ideal inductor's average power. (b) DC through an ideal inductor after transients yields zero voltage drop and zero power. (e) Invalid since a correct statement exists. Common Pitfalls: Confusing reactive power with real power; assuming frequency dependence of current equates to power loss; ignoring non-ideal resistive effects. Final Answer: There is no meaningful relationship (ideal inductor has zero average power at any frequency).

Question 2

Power from voltage and resistance — corrected stem (Recovery-First): Assume the component is a 1 kΩ resistor (not “1000 W”). With 25 V applied across it, how much power does it dissipate?

Accepted Answer

Introduction / Context: The original text likely intended “1000 Ω resistor,” not “1000 W resistance.” Using the Recovery-First policy, we minimally correct the stem to a realistic component value. The problem then becomes a straightforward application of power formulas for resistive elements. Given Data / Assumptions: Resistance R = 1 kΩ = 1000 Ω. Applied voltage V = 25 V (DC or RMS AC). Ideal resistor; temperature and tolerance effects neglected. Concept / Approach: For a resistor, power can be computed as P = V^2 / R, equivalently P = I^2 * R or P = V * I. With V and R given, the V^2 / R form is the most direct. Always check units: volts squared over ohms yields watts. Step-by-Step Solution: Compute V^2: 25 * 25 = 625.Divide by R: 625 / 1000 = 0.625.Thus, P = 0.625 W. Verification / Alternative check: Find current first: I = V / R = 25 / 1000 = 0.025 A (25 mA). Then P = V * I = 25 * 0.025 = 0.625 W. The two methods agree. Why Other Options Are Wrong: (a) 0.24 W results from incorrect arithmetic. (c) 80 W would require much lower resistance or higher voltage. (d) 325 kW is wildly unrealistic in this context. (e) Invalid because a correct option exists. Common Pitfalls: Misreading “1000 W” for watts instead of ohms; forgetting unit conversions; squaring voltage but not dividing by resistance. Final Answer: 0.625 W.

Question 3

Switch fundamentals — normally open (NO) push button: Under what condition can current flow through a normally open push button switch in a simple series circuit?

Accepted Answer

Introduction / Context: Understanding switch contact states is essential for troubleshooting and designing basic control circuits. A normally open (NO) push button is used for momentary activation—think doorbells, start buttons, and user inputs to logic systems. Given Data / Assumptions: Switch type: normally open (NO) push button. Simple series circuit with a power source and a load. Ideal contacts: open = infinite resistance (no current), closed = low resistance (current allowed). Concept / Approach: By definition, an NO contact remains open in its rest state. Pressing the actuator mechanically closes the contacts, completing the circuit path so current can flow. Releasing the button opens the contacts again, breaking the circuit and stopping current. Step-by-Step Solution: Rest state: NO contact is open → circuit open → current = 0.Press button: contacts close → circuit completed → load current flows.Release button: contacts open → circuit broken → current stops. Verification / Alternative check: Compare with a normally closed (NC) push button: NC passes current at rest and interrupts when pressed. This contrast validates the NO behavior described. Why Other Options Are Wrong: (b) and (c) state “open” permits current; incorrect. (d) Reasserts the “open” condition and remains false even with a supply. (e) Invalid because a correct choice exists. Common Pitfalls: Confusing NO with NC; assuming indicator lights or control logic invert the meaning; overlooking wiring errors that can mask intended behavior. Final Answer: Only while you are pressing the button (contacts closed).

Question 4

DC current gain computation (BJT): A transistor has a current gain (β) of 175. If the base current is 0.1 mA, calculate the collector current.

Accepted Answer

Introduction / Context: For a bipolar junction transistor (BJT) operating in the active region, the DC current gain β (also written h_FE) relates collector current to base current as I_C = β * I_B. This quick calculation is routine when sizing bias networks, estimating load current capability, or checking feasibility. Given Data / Assumptions: β = 175 (dimensionless DC current gain). I_B = 0.1 mA. Transistor assumed to be in forward-active operation (not saturated). Concept / Approach: Use the definition of DC current gain: I_C = β * I_B. Ensure units are consistent. Convert milliampere if desired, but here the numbers multiply cleanly without conversion as long as we keep track of the decimal places. Step-by-Step Solution: Write formula: I_C = β * I_B.Substitute: I_C = 175 * 0.1 mA.Compute: 175 * 0.1 = 17.5.Therefore, I_C = 17.5 mA. Verification / Alternative check: Convert base current to amperes: 0.1 mA = 0.0001 A. Then I_C = 175 * 0.0001 A = 0.0175 A = 17.5 mA. Same result. Why Other Options Are Wrong: (a) 1750 mA is 100 times too large (as if β = 1750 or I_B misread). (c) 175 mA is 10 times too large. (d) 1.75 mA is 10 times too small. (e) Invalid since a correct answer exists. Common Pitfalls: Confusing β with gain in percent; misplacing the decimal with milliampere units; applying the formula when the device is saturated (where the β relation breaks down). Final Answer: 17.5 mA.

Question 5

Intrinsic semiconductors at room temperature: An intrinsic (undoped) semiconductor has some holes present. What primarily causes the creation of these holes?

Accepted Answer

Introduction / Context: In an intrinsic semiconductor such as pure silicon, charge carriers (electrons and holes) exist due to thermal excitation. Understanding the origin of intrinsic carriers is fundamental to band theory, diode operation, and temperature dependence of conductivity. Given Data / Assumptions: Material is intrinsic (undoped). Room temperature conditions (thermal energy available). We seek the mechanism that creates holes without impurities. Concept / Approach: Thermal energy excites some valence electrons across the bandgap into the conduction band. When an electron leaves the valence band, it leaves behind a vacant state—called a hole. In intrinsic material, electrons and holes are generated in pairs, so n = p, and both concentrations rise with temperature according to an exponential dependence on the bandgap. Step-by-Step Solution: Start with a fully occupied valence band at absolute zero.At room temperature, thermal energy occasionally promotes electrons to the conduction band.Each promoted electron leaves a hole behind in the valence band.Thus, intrinsic carriers (electrons and holes) arise due to thermal generation-recombination equilibrium. Verification / Alternative check: Carrier concentration in intrinsic silicon n_i increases with temperature. No dopants are needed; adding dopants would make the material extrinsic and unbalance n and p. Why Other Options Are Wrong: (a) Doping intentionally adds impurities; intrinsic means no dopants. (b) “Valence electrons” exist by definition, but their mere presence does not create holes without excitation. (c) Free electrons are the counterpart carriers created together with holes; they are an effect, not the cause. Common Pitfalls: Confusing intrinsic and extrinsic semiconductors; assuming holes are “preexisting” rather than generated with electrons; overlooking the temperature dependence. Final Answer: Thermal energy.

Question 6

Parallel R–L–C network behavior: When operating above its resonant frequency, a parallel RLC circuit most closely behaves like which simpler parallel network?

Accepted Answer

Introduction / Context: A parallel RLC circuit has a resistor, inductor, and capacitor connected across the same AC source. Its overall behavior depends on frequency. Below, at, and above the resonant frequency, the net reactive effect (inductive or capacitive) changes. Knowing which tendency dominates above resonance helps in filter design and tuning. Given Data / Assumptions: Ideal sinusoidal steady state. Elements R, L, and C are in parallel. Resonant frequency f0 where total susceptance is zero. Concept / Approach: For a parallel network, work with admittances: Y = G + jB. The susceptances are B_C = +ωC and B_L = −1/(ωL). At resonance, B_C + B_L = 0. When frequency increases above f0, B_C increases in magnitude (directly proportional to ω), while |B_L| decreases (inversely proportional to ω). The sum B = B_C + B_L becomes positive, meaning net capacitive behavior. Hence the circuit behaves like a parallel RC network as a first-order approximation. Step-by-Step Solution: Write susceptances: B_C = ωC, B_L = −1/(ωL).At f0: ω0C = 1/(ω0L) so B_C + B_L = 0.For f > f0: ω increases ⇒ B_C increases; 1/(ωL) decreases ⇒ |B_L| shrinks.Thus B = B_C + B_L > 0 ⇒ net capacitive.Therefore, the circuit resembles a parallel RC. Verification / Alternative check: Phasor diagram of branch currents shows the capacitive branch current dominating above f0, pulling total current to lead the voltage, which is the hallmark of capacitive behavior. Why Other Options Are Wrong: (a) Net inductive behavior occurs below resonance, not above. (c) While still an RLC, simplification to an equivalent RC captures the dominant effect. (d) Purely resistive occurs only at exact resonance (ideal case), not above it. Common Pitfalls: Confusing series and parallel resonance trends; assuming the same lead/lag rules apply in both topologies without checking susceptance signs. Final Answer: Parallel RC circuit (net capacitive behavior)

Question 7

Series-aiding sources (DC addition): Two ideal voltage sources of 6 V and 9 V are connected in series aiding. What is the total source voltage supplied to the circuit?

Accepted Answer

Introduction / Context: When DC sources are connected in series aiding, their voltages add algebraically because the same current flows through both and their polarities support each other. This is common in battery packs and laboratory supplies arranged in series to reach a higher voltage. Given Data / Assumptions: Two ideal sources: 6 V and 9 V. Series aiding polarity (positive of one tied to negative of the other so voltages add). Neglect internal resistance for simplicity. Concept / Approach: Series-aiding voltages add directly: V_total = V1 + V2. If the sources were opposing, the difference would apply instead. Here, aiding is explicit, so addition is appropriate. Step-by-Step Solution: Identify polarity as aiding.Compute V_total = 6 V + 9 V.V_total = 15 V. Verification / Alternative check: Draw a simple loop and apply KVL: the sum of source rises equals the net rise, confirming 15 V available to the load. Why Other Options Are Wrong: (b) 16 V would require 7 V and 9 V or an error. (c) 7.5 V is an average, not applicable. (d) 8 V is unrelated. (e) Not needed as an exact value is known. Common Pitfalls: Mixing aiding with opposing; forgetting that source internal resistances, if present, affect current but not the ideal sum of voltages. Final Answer: 15 V

Question 8

Zener regulator (no-load): An unloaded zener regulator has source voltage 20 V, series resistor 330 Ω, and zener voltage 12 V. What is the zener current under these conditions?

Accepted Answer

Introduction / Context: A basic shunt zener regulator uses a series resistor to drop excess voltage so the zener diode can clamp the output near its zener voltage. With no load attached, nearly all current through the series resistor flows through the zener. Computing this current is a straightforward application of Ohm’s law. Given Data / Assumptions: Source voltage V_S = 20 V. Zener voltage V_Z = 12 V (assume ideal regulation at this value). Series resistor R_S = 330 Ω. No-load (unloaded) condition: I_S ≈ I_Z. Concept / Approach: Series resistor current I_S = (V_S − V_Z) / R_S. With no load, the same current becomes the zener current I_Z. This ensures the zener remains in regulation as long as I_Z stays within its allowable range. Step-by-Step Solution: Compute voltage across R_S: V_RS = V_S − V_Z = 20 − 12 = 8 V.Compute I_S: I_S = V_RS / R_S = 8 / 330 A.I_S ≈ 0.02424 A = 24.24 mA.Therefore I_Z ≈ 24.2 mA (no-load). Verification / Alternative check: Sanity check: If R_S were 400 Ω, I_Z would be 20 mA; with 330 Ω, a bit higher than 20 mA is expected, consistent with 24.2 mA. Why Other Options Are Wrong: (a) and (c) overstate the current for the given resistor. (d) is far too high and would require a much smaller resistor. Common Pitfalls: Forgetting to subtract the zener voltage from the source; mixing mA and A; ignoring load conditions which, if present, would reduce I_Z accordingly. Final Answer: 24.2 mA

Question 9

Purely capacitive AC circuit: If the operating frequency is doubled while the RMS voltage is held constant, how does the total circuit current change?

Accepted Answer

Introduction / Context: In a purely capacitive AC circuit, the reactance depends on frequency. This determines the current drawn for a fixed applied voltage. Understanding this relationship is fundamental for capacitor sizing, reactive power considerations, and filter behavior. Given Data / Assumptions: Ideal capacitor C with negligible series resistance. RMS voltage V is constant. Frequency is changed from f to 2f. Concept / Approach: Capacitive reactance is X_C = 1/(2πfC). The current in a pure capacitor is I = V / X_C = V * 2πfC. Therefore I is directly proportional to f when V and C are constant. Doubling f doubles I. Step-by-Step Solution: Start: X_C = 1/(2πfC).Current: I = V / X_C = V * 2πfC.If f → 2f, then I → 2I.Conclusion: current doubles. Verification / Alternative check: Example numbers: Let V = 10 V, C = 100 μF, f = 50 Hz. I1 = 10 * 2π * 50 * 100e−6 ≈ 0.314 A. At 100 Hz, I2 ≈ 0.628 A, exactly double. Why Other Options Are Wrong: (a) ignores the frequency dependence. (c) and (d) would require inverse or triple proportionality, which is not the case. (e) is unnecessary because a precise rule exists. Common Pitfalls: Using X_C ∝ 1/f but then forgetting that I = V / X_C ∝ f; mixing peak and RMS values does not change the proportionality. Final Answer: Current doubles

Question 10

Parallel combination magnitude: A 100 Ω resistor is connected in parallel with a capacitor whose reactance magnitude is 100 Ω at the operating frequency. What is the magnitude of the total impedance |Z|?

Accepted Answer

Introduction / Context: When resistive and reactive branches are in parallel, it is usually easier to work with admittances and then invert to find the impedance magnitude. This problem is a classic symmetric case where the resistor magnitude equals the reactance magnitude, simplifying the arithmetic. Given Data / Assumptions: Resistor R = 100 Ω. Capacitive reactance magnitude |X_C| = 100 Ω (so branch impedance is −j100 Ω). Ideal components, single frequency. Concept / Approach: Admittance of the network is Y = 1/R + 1/(−jX_C) = 1/R + j(1/X_C). The magnitude is |Y| = sqrt((1/R)^2 + (1/X_C)^2). Then |Z| = 1/|Y|. With R = X_C = 100 Ω, the symmetry yields a simple numerical result. Step-by-Step Solution: Compute conductance: G = 1/R = 1/100 = 0.01 S.Compute susceptance: B = 1/X_C = 1/100 = 0.01 S.Admittance magnitude: |Y| = sqrt(G^2 + B^2) = sqrt(0.01^2 + 0.01^2) = 0.014142 S.Impedance magnitude: |Z| = 1/|Y| ≈ 1/0.014142 ≈ 70.7 Ω. Verification / Alternative check: A numeric phasor or calculator check yields the same 70.7 Ω. This is consistent with the general relation for equal R and X branches in parallel: |Z| = R/√2. Why Other Options Are Wrong: (a) 60 Ω is too low; (b) ~99 Ω suggests near-resistive behavior, not correct here; (c) 250 Ω is far off. The computed value matches 70.7 Ω. Common Pitfalls: Adding impedances directly in parallel instead of using admittance; forgetting to take magnitude after combining real and imaginary parts. Final Answer: 70.7 Ω

Question 11

Norton–Thevenin conversion: A circuit has Norton current I_N = 10 mA and Norton resistance R_N = 10 kΩ. What is the equivalent Thevenin source?

Accepted Answer

Introduction / Context: Norton and Thevenin equivalents are dual representations: Norton uses a current source in parallel with a resistance, while Thevenin uses a voltage source in series with a resistance. Converting between them is routine in network analysis and simplifies load calculations. Given Data / Assumptions: Norton current I_N = 10 mA. Norton resistance R_N = 10 kΩ. Ideal linear network, so R_N = R_TH. Concept / Approach: The conversion uses V_TH = I_N * R_N and R_TH = R_N. The topology changes from current source in parallel to voltage source in series, preserving terminal behavior for any load. Step-by-Step Solution: Compute V_TH = I_N * R_N = 0.01 A * 10,000 Ω = 100 V.Set R_TH = R_N = 10 kΩ.Therefore Thevenin equivalent is 100 V in series with 10 kΩ. Verification / Alternative check: Open-circuit voltage equals 100 V, and short-circuit current equals 10 mA in both forms, confirming equivalence. Why Other Options Are Wrong: (a) 1000 V is a decade too high. (b) and (d) incorrectly keep the resistance in parallel in the Thevenin form. Common Pitfalls: Mixing series/parallel placement of the resistance; miscomputing V_TH by forgetting to convert mA to A. Final Answer: 100 V with series resistance 10 kΩ

Question 12

Inductor geometry effect: If the diameter of a coil in an inductor is doubled (other factors like turns and length held constant), how does the inductance change?

Accepted Answer

Introduction / Context: Inductance depends on geometry as well as core material. For common air-core solenoids, inductance is roughly proportional to N^2 * A / l, where N is turns, A is cross-sectional area, and l is length. Understanding how size changes alter L is crucial for RF coils and power inductors. Given Data / Assumptions: Coil diameter is doubled. Number of turns N and coil length l are unchanged. Core properties unchanged. Concept / Approach: Area A of the coil cross-section scales with the square of diameter (A ∝ d^2). With L ∝ N^2 * A / l and N, l constant, L ∝ A ∝ d^2. Doubling diameter (d → 2d) makes area four times larger, so inductance increases by a factor of four. Step-by-Step Solution: Initial relation: L ∝ N^2 * A / l.A ∝ d^2; N, l fixed.If d → 2d, then A → 4A.Therefore L → 4L (inductance quadruples). Verification / Alternative check: Empirical inductor calculators show L scaling roughly with d^2 for a fixed-length, fixed-turn solenoid, aligning with the proportionality argument. Why Other Options Are Wrong: (a) only doubles instead of quadruples; (b) and (c) wrongly suggest a decrease in L with larger diameter. Common Pitfalls: Forgetting that area grows with diameter squared; confusing changes in diameter with changes in the number of turns. Final Answer: Inductance is multiplied by four

Question 13

Step-down transformer facts: Which statement correctly describes a step-down power transformer (ideal case, ignoring losses)?

Accepted Answer

Introduction / Context: A step-down transformer reduces voltage from primary to secondary while ideally conserving power (P ≈ constant, ignoring losses). Consequently, when voltage goes down, current must increase proportionally to maintain the same power. Recognizing this trade-off is central to transformer-based power conversion. Given Data / Assumptions: Ideal transformer relationships apply. Step-down implies V_s < V_p (secondary voltage less than primary). Power conservation approximation: V_p * I_p ≈ V_s * I_s. Concept / Approach: For an ideal transformer: V_s/V_p = N_s/N_p and I_s/I_p = N_p/N_s. If V_s is decreased (N_s < N_p), then I_s increases relative to I_p, assuming the same load power transfer. Thus, step-down in voltage implies step-up in current. Step-by-Step Solution: Recognize step-down: V_s < V_p.Apply ideal relation: I_s/I_p = V_p/V_s > 1.Therefore I_s > I_p.Conclude the correct statement: secondary current exceeds primary current. Verification / Alternative check: Example: If V_p = 120 V, V_s = 12 V (10:1). Delivering 120 W implies I_p = 1 A and I_s = 10 A, confirming the current increase on the secondary. Why Other Options Are Wrong: (a) contradicts step-down definition. (c) and (d) are unrelated statements about different circuits and are not transformer properties. Common Pitfalls: Confusing voltage and current ratios; forgetting that power is approximately conserved (minus losses), so as voltage drops, current rises. Final Answer: Secondary current is greater than primary current (for the same power level)

Question 14

Energy conversion devices: Which device converts chemical energy directly into electrical energy in practical electrical systems?

Accepted Answer

Introduction / Context: Different transducers and sources convert one form of energy into another. Identifying which device accomplishes a specific conversion is foundational knowledge for electrical engineering and physics students. Here the focus is on chemical-to-electrical conversion, which powers most portable electronics. Given Data / Assumptions: Common devices: solar cells, thermocouples, batteries, motors. Direct energy conversion is considered (no intermediate mechanical stage). Practical, widely used implementations. Concept / Approach: A chemical cell (battery) uses electrochemical reactions at electrodes separated by an electrolyte to generate an electromotive force. This EMF drives electrons through an external circuit, delivering electrical energy. By contrast, solar cells convert light to electricity (photovoltaic effect); thermocouples convert temperature differences to electricity (Seebeck effect); motors convert electrical to mechanical energy. Step-by-Step Solution: Identify the requested conversion: chemical → electrical.Match to device: electrochemical cell (battery) performs this conversion.Exclude others: solar (light → electrical), thermocouple (thermal → electrical), motor (electrical → mechanical). Verification / Alternative check: Everyday examples include alkaline, lithium-ion, and lead–acid batteries; all provide electrical power from chemical reactions. Why Other Options Are Wrong: (a) Solar cell uses photons, not chemical reactions. (b) Thermocouple uses a temperature gradient. (d) Motor converts in the opposite direction. Common Pitfalls: Confusing “fuel cell” and “battery”; both are chemical-to-electrical, and both fall under the broader category of chemical cells. Final Answer: Chemical cell (battery)

Question 15

Magnetic saturation in an electromagnet core: When is a ferromagnetic core said to be in saturation during coil excitation?

Accepted Answer

Introduction / Context: Magnetic saturation is a material property of ferromagnetic cores used in inductors and transformers. It limits how much magnetic flux can be increased by raising the magnetizing force (mmf = N * I). Recognizing saturation is vital for avoiding distortion, excessive magnetizing current, and core heating. Given Data / Assumptions: Ferromagnetic core within a coil. Magnetizing force increased by increasing current. Idealized focus on B–H relationship. Concept / Approach: In the B–H curve of ferromagnetic materials, the slope dB/dH (permeability) is high at low H, then flattens as the material approaches saturation. Past this knee, large increases in current (and thus H) yield only small increases in flux density B. This operational region is called saturation and is undesirable in linear applications. Step-by-Step Solution: Relate mmf to current: H ∝ N * I / l.Observe B–H: initially steep, then flattens near saturation.Define saturation: further current increases produce little increase in B.Therefore, saturation corresponds to option describing minimal B change for added I. Verification / Alternative check: Measure magnetizing current versus flux: near saturation, current rises disproportionately with only small flux gains, often seen as waveform distortion and overheating in transformers. Why Other Options Are Wrong: (a) Mechanical movement is unrelated. (c) Heating may occur but is not the definition. (d) Field polarity reversal with current reversal happens at all times and does not indicate saturation. Common Pitfalls: Confusing mmf with flux density; mmf (N*I) always increases with current, but flux density B does not increase proportionally once saturation is approached. Final Answer: When increasing coil current produces little to no further increase in magnetic flux density

Question 16

In AC circuit theory, consider an ideal (pure) capacitor with no series resistance: what is the correct statement about its average real power dissipation as operating frequency changes?

Accepted Answer

Introduction / Context: When studying AC circuits, it is essential to distinguish between energy storage in reactive components and true (average) power dissipation. An ideal capacitor stores energy in an electric field and exchanges this energy with the source each cycle. This question tests your understanding of why a pure capacitor does not dissipate average real power and why this conclusion holds regardless of frequency, provided the capacitor is ideal (no resistance). Given Data / Assumptions: The component is an ideal capacitor with zero series resistance and zero dielectric loss (no ESR, no leakage). The source is sinusoidal with any frequency f. Average real power refers to the time-average over one or more cycles of v(t) * i(t). Concept / Approach: For an ideal capacitor, current leads voltage by 90 degrees. With i(t) = C * dv/dt for a sinusoid, the instantaneous power p(t) = v(t) * i(t) alternates positive and negative. The energy taken from the source in part of the cycle is returned in another part. The average over a complete cycle is zero. This is true at all frequencies; frequency changes the magnitude of reactive current (I = 2 * pi * f * C * V_rms) and reactive power, but not the average real power of an ideal capacitor. Step-by-Step Solution: Note the phase relationship: current leads voltage by 90 degrees in a pure capacitor.Write p(t) = v(t) * i(t). Positive and negative lobes are equal and opposite over a cycle.Average real power P_avg = (1/T) * ∫ p(t) dt over one period = 0.Since the ideal model is frequency-independent for losses, P_avg = 0 for any f. Verification / Alternative check: Compute with phasors: P = V_rms * I_rms * cos(phi). For an ideal capacitor, phi = −90 degrees, so cos(phi) = 0 and P = 0 at all frequencies. Only reactive power Q = V_rms * I_rms * sin(phi) is nonzero in magnitude. Why Other Options Are Wrong: Power increasing or decreasing with frequency (B or C) describes nonideal ESR loss, not a pure capacitor. “No definable relationship” (D) is incorrect because the relationship is precisely defined: average real power is zero. “None” is invalid because the correct statement is given in option A. Common Pitfalls: Confusing reactive current growth with real power dissipation; forgetting ESR and dielectric losses belong to nonideal models only; assuming “pulsating power” implies nonzero average—here the average is zero. Final Answer: Average real power dissipation is zero and does not depend on frequency

Question 17

In bipolar junction transistors (BJTs), how does the DC current gain (beta, β) generally vary as junction temperature increases, under typical operating conditions?

Accepted Answer

Introduction / Context: The DC current gain β of a bipolar junction transistor is not a fixed constant. It depends on device geometry, collector current, and temperature. Recognizing the qualitative temperature trend of β is important for bias stability, thermal design, and predicting the shift of operating points in amplifiers and switches. Given Data / Assumptions: The device is a silicon BJT operated in the forward-active region. Temperature rises within normal operating limits (no breakdown or thermal runaway). We are concerned with the typical trend, not extreme edge cases. Concept / Approach: As temperature increases, carrier mobility and recombination mechanisms change. For most silicon BJTs in the common operating range, β tends to increase with temperature because the base-emitter voltage required for a given collector current decreases, and minority carrier lifetimes change in a way that effectively boosts the collector current for a fixed base drive. Designers exploit or guard against this by using negative feedback, emitter resistors, or bias-stabilizing networks to maintain a stable Q point. Step-by-Step Solution: Identify the parameter: β = I_C / I_B in forward-active region.Note temperature effect: V_BE falls roughly 2 mV/°C; recombination dynamics shift.Result: For a fixed base current, I_C typically increases with temperature → β increases.Choose “increases.” Verification / Alternative check: Examine datasheets: β (h_FE) curves often show higher gain at elevated junction temperatures for a given I_C, up to a region where excessive current or high temperature can reverse the trend or induce breakdown—outside normal design targets. Why Other Options Are Wrong: “decreases” contradicts the common midrange trend; “constant” ignores real temperature dependence; “can be any of the above” is too vague for typical design assumptions; “None” is incorrect because a clear trend exists in standard usage. Common Pitfalls: Assuming β is constant; overlooking that β also varies with I_C; forgetting to include temperature compensation in precision analog designs. Final Answer: increases

Question 18

Transistor biasing and load-line analysis: if the base supply (bias) voltage increases in a fixed-bias amplifier, how does the operating point (Q point) move on the DC load line?

Accepted Answer

Introduction / Context: The quiescent point (Q point) defines the steady-state collector current and collector-emitter voltage in a BJT amplifier. Understanding how the Q point shifts with bias changes is key for setting linear operating regions, avoiding cutoff or saturation, and ensuring headroom for signal swings. This question focuses on how increasing base bias affects the Q point along a fixed load line determined by R_C and V_CC. Given Data / Assumptions: Common-emitter amplifier with a fixed collector resistor R_C and supply V_CC. Increasing base supply (bias) voltage increases base current I_B. Transistor remains in forward-active region initially. Concept / Approach: With greater base drive, collector current I_C = β * I_B rises. On the DC load line (a straight line from (V_CE = V_CC, I_C = 0) to (V_CE = 0, I_C = V_CC/R_C)), increasing I_C moves the operating point upward (toward higher I_C) and leftward (toward lower V_CE). The conventional phrasing “moves up” refers to the vertical axis (collector current). Step-by-Step Solution: Increase base bias → I_B increases.I_C = β * I_B increases, assuming β roughly constant locally.On the load line, higher I_C corresponds to a point higher up the vertical axis.Therefore the Q point moves up (and left) along the load line. Verification / Alternative check: Plotting I_C vs V_CE under different I_B curves shows intersection with the fixed load line migrating upward as I_B increases; oscilloscopes and curve tracers demonstrate the same trend experimentally. Why Other Options Are Wrong: “down” is opposite to the effect of increased I_B. “nowhere” implies no change, which is incorrect. “off the load line” would require changing the circuit or violating KVL; bias changes simply shift the operating point along the same load line until saturation or cutoff. Common Pitfalls: Ignoring β variation with current; forgetting that extreme increases can push the transistor into saturation where load-line linear analysis no longer applies. Final Answer: up

Question 19

Household mains example: if the outlet provides 120 V rms (root-mean-square), what is the corresponding peak voltage of the sinusoidal waveform?

Accepted Answer

Introduction / Context: AC specifications often use RMS values because they directly relate to equivalent DC heating effect in resistive loads. However, many design calculations—such as capacitor voltage ratings or rectifier sizing—require the peak voltage. Converting between RMS and peak for a pure sine wave is a basic skill in power electronics and instrumentation. Given Data / Assumptions: Waveform is a sinusoid. RMS voltage V_rms = 120 V. Standard relationship for sine waves applies: V_peak = √2 * V_rms. Concept / Approach: For a sinusoidal signal, V_peak = V_rms * sqrt(2), and V_pp (peak-to-peak) = 2 * V_peak. These arise from the RMS definition V_rms = V_peak / sqrt(2). The conversion is exact for a sine wave and independent of frequency. Step-by-Step Solution: Use V_peak = sqrt(2) * V_rms.Compute: V_peak = 1.4142... * 120 ≈ 169.7 V.Round sensibly to 170 V for component selection and reporting.Select 170 V. Verification / Alternative check: Reverse the calculation: V_rms = V_peak / sqrt(2) = 170 / 1.414 ≈ 120 V, confirming consistency. Why Other Options Are Wrong: 100 V and 120 V are below the true peak; 180 V exceeds the precise value and is not the standard conversion. “None” is invalid because 170 V is correct. Common Pitfalls: Confusing peak with peak-to-peak (which would be ≈ 340 V); using approximate factors like 1.5 instead of sqrt(2); forgetting that non-sinusoidal waveforms require a different approach. Final Answer: 170 V

Question 20

In BJT operation, which bias condition is typically applied to the emitter-base junction (the “emitter diode”) during normal amplifier operation?

Accepted Answer

Introduction / Context: A bipolar junction transistor has two pn junctions: emitter-base (EB) and collector-base (CB). Understanding their biasing in normal operation clarifies current flow, gain mechanisms, and why small base currents control large collector currents. In amplifier mode (forward-active region), the EB junction must promote carrier injection from the emitter into the base. Given Data / Assumptions: Transistor used as a linear amplifier (forward-active region). Proper supply voltages provided. No saturation or breakdown intended. Concept / Approach: In forward-active operation, the emitter-base junction is forward-biased to inject majority carriers from the emitter into the base. The collector-base junction is reverse-biased to sweep these carriers into the collector region. This arrangement yields current amplification characterized by β = I_C / I_B. Reversing these biases moves the device into cutoff, saturation, or reverse active modes, none of which correspond to standard small-signal amplification. Step-by-Step Solution: Identify normal amplifier mode: forward-active.Recall EB junction condition: forward-biased to allow injection.Recall CB junction condition: reverse-biased for collection.Choose “Forward-biased” for the EB junction. Verification / Alternative check: Examine transistor characteristic curves: with EB forward and CB reverse, output characteristics show controlled collector current; datasheets and textbooks depict this mode explicitly. Why Other Options Are Wrong: Reverse-biased EB junction would cut off emitter injection. “Nonconducting” conflicts with amplifier operation. “Breakdown region” is undesirable and destructive for normal operation. “None” is incorrect because forward bias is required. Common Pitfalls: Confusing saturation (both junctions forward-biased) with forward-active; overlooking that small V_BE changes produce large I_C variations, demanding bias stabilization. Final Answer: Forward-biased

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