Q: RLC damping change: A series RLC circuit is underdamped. To convert its response to overdamped, how must the resistance R be adjusted while keeping L and C fixed?

Introduction / Context: The transient response of a second-order series RLC circuit (to steps or impulses) depends on the damping ratio ζ, which is controlled by resistance R for fixed inductance L and capacitance C. This question asks how to move from underdamped (oscillatory) to overdamped (non-oscillatory) behavior by adjusting R. Given Data / Assumptions: Series RLC with fixed L and C; only R is varied. Underdamped initially, so ζ 1. Concept / Approach: For a series RLC, the characteristic equation is s^2 + (R/L)s + 1/(LC) = 0. The damping ratio is ζ = (R/2) * sqrt(C/L) in normalized units. The regimes are: underdamped if R 2√(L/C). Thus, increasing R beyond the critical value transitions the system through critical to overdamped behavior. Step-by-Step Solution: Start condition: underdamped ⇒ R 2√(L/C).Therefore, R must be increased (past the critical value). Verification / Alternative check: Compare time responses: increasing R reduces oscillation amplitude and eventually eliminates oscillation when ζ ≥ 1, confirming the qualitative effect of higher resistance. Why Other Options Are Wrong: Decrease R: Reduces damping → more oscillatory (further underdamped). Increase to infinity: Open circuit removes dynamics rather than producing a practical overdamped response. Reduce to zero: Yields an undamped LC oscillator (ζ = 0). Common Pitfalls: Mixing up parallel vs. series RLC damping conditions; formulas differ but the trend—larger R in series increases damping—holds. Final Answer: has to be increased

Question 1

Current continuity in conductors: When a steady current flows through a conductor with a non-uniform cross-sectional area, how do current and current density vary along different cross-sections?

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Introduction / Context: In electrical engineering and physics, understanding how electric current behaves in conductors with changing geometry is fundamental. This question tests current continuity and the relationship between current (I), current density (J), and cross-sectional area (A) for a steady (time-invariant) current in a single connected conductor. Given Data / Assumptions: Direct current or steady-state current (no charge accumulation with time). Conductor is continuous, with no branches and no charge sinks/sources along its length. Area of cross-section varies along the length (non-uniform A). Concept / Approach: Conservation of charge (continuity) requires the same current to flow through every cross-section in steady state. However, current density is defined as J = I / A in a uniform material when current distributes evenly across the section. If A changes, J must adjust inversely to maintain the same I. Hence, I is constant, while J varies with A. Step-by-Step Solution: Let I be the steady current entering the conductor.At cross-section 1 with area A1: J1 = I / A1.At cross-section 2 with area A2: J2 = I / A2.Since there are no nodes/branches, I1 = I2 = I (current continuity).If A2 < A1, then J2 > J1, and vice versa. Verification / Alternative check: Use the integral form of the continuity equation for steady state: ∇·J = 0. For a prismatic path with varying A(s), integrating over a pillbox aligned with the conductor gives equal inflow and outflow currents, confirming I is constant while J redistributes with A. Why Other Options Are Wrong: Option (a): Different I at different sections violates charge conservation in steady state. Option (b): Implies constant I but says nothing about J; incomplete and not the best answer to the combined assertion. Option (c): Suggests constant J but varying I; physically inconsistent for a single connected path. Option (e): Both I and J cannot remain the same when A changes; J must vary as 1/A. Common Pitfalls: Confusing current with current density; current is global through the section, current density is local and area-dependent. Assuming branching or storage; neither applies in a single, steady path. Final Answer: current will be the same but current density will be different at different cross-sections.

Question 2

Phase relation in an R–C series circuit: For R = 100 Ω in series with a capacitor having reactance Xc = 10 Ω, what is the approximate phase angle between current and applied voltage?

Accepted Answer

Introduction / Context: Series resistor–capacitor (R–C) circuits exhibit a characteristic phase shift between current and voltage due to the reactive nature of the capacitor. This question assesses your ability to compute the phase angle using given resistance and capacitive reactance values and to interpret whether current leads or lags the applied voltage. Given Data / Assumptions: R = 100 Ω (resistance). Xc = 10 Ω (capacitive reactance at the operating frequency). Sinusoidal steady-state conditions. Concept / Approach: For a series R–C circuit, the impedance is Z = R − j Xc. The current leads the applied voltage by an angle θ where tan θ = Xc / R (positive lead for capacitive circuits). The magnitude of θ is small when Xc ≪ R and large when Xc ≫ R. Step-by-Step Solution: Compute the phase angle: θ = arctan(Xc / R).Substitute values: θ = arctan(10 / 100) = arctan(0.1).Numerically, arctan(0.1) ≈ 5.71°, which is approximately 6°.Because the reactance is capacitive, the current leads the voltage by θ. Verification / Alternative check: Phasor diagram: current I taken as reference; voltage across R is in phase with I, voltage across C lags I by 90°. The resultant applied voltage lags I by θ, so I leads V by θ. With R ≫ Xc, the angle is indeed small (≈ 6°). Why Other Options Are Wrong: In phase: only true when Xc = 0 (purely resistive). Lead by 84°: would require Xc ≫ R, not the case here. Lagging options: incorrect sign; capacitive circuits cause current to lead voltage. Common Pitfalls: Using θ = arctan(R / Xc), which is the complement and gives the wrong magnitude. Confusing lead/lag conventions between current and voltage. Final Answer: the current leads the voltage by about 6°

Question 3

Cable ampacity basics: The current-carrying rating of an insulated power cable primarily depends on which physical parameter(s)?

Accepted Answer

Introduction / Context: Cable current rating (ampacity) is the maximum continuous current a cable can carry without exceeding its temperature limit. This rating is governed by heat generation (I^2 * R) and heat dissipation to the surroundings through the insulation and sheath. The question probes which geometric property most directly affects ampacity in basic terms. Given Data / Assumptions: Comparing fundamental geometric effects (length vs. diameter) on current rating. Installation and environmental conditions are considered fixed. Steady-state thermal balance (no transients). Concept / Approach: For a given material and insulation system, conductor cross-sectional area (which scales with diameter for round conductors) determines resistance per unit length and the heat generated at a given current. Larger diameter (greater area) lowers resistance, reduces I^2 * R losses per unit length, and increases permissible current before reaching the temperature limit. Cable length does not directly change ampacity; it changes total voltage drop and total heat but not the allowable current per unit length that meets temperature constraints under uniform conditions. Step-by-Step Solution: Relate resistance per unit length r ∝ 1/Area; Area ∝ diameter^2.Heat generation per unit length q = I^2 * r → decreases with larger diameter.For fixed thermal environment, higher diameter permits higher I before insulation limit is reached.Length affects total drop (ΔV = I * r * L) but not the per-unit-length thermal limit that defines ampacity. Verification / Alternative check: Thermal circuit models show conductor temperature rise ΔT ∝ (I^2 * r) / G_th, where G_th is thermal conductance to ambient per unit length. Increasing diameter reduces r and usually increases G_th slightly, improving rating. Why Other Options Are Wrong: Length: does not set the allowable current; it affects voltage drop and total heating, not the fundamental ampacity. Both length and diameter: overstates the role of length in defining rating. None/ambient only: geometry (diameter/area) is essential. Common Pitfalls: Confusing voltage-drop limits (length matters) with thermal ampacity (dominated by cross-section and installation conditions). Final Answer: diameter of cable

Question 4

Inductance unit equivalence: One henry (H) is equal to which of the following unit ratios?

Accepted Answer

Introduction / Context: Recognizing equivalent forms of SI units helps in dimensional analysis and understanding physical definitions. The henry (H) is the SI unit of inductance used widely in circuit theory, electromagnetics, and power systems. Given Data / Assumptions: Standard SI base and derived units. Definition of inductance from magnetic flux–current relation or voltage–current dynamics. Concept / Approach: Inductance L is defined by the relations: flux linkage λ = L * I and v = L * di/dt for a linear inductor. From λ = N * Φ (for N turns) and L = λ / I, the unit of L is weber-turns per ampere; for one-turn, it reduces to weber per ampere. The voltage form v = L di/dt implies L has units of volt-second per ampere, which equals weber per ampere because 1 weber = 1 volt·second. Step-by-Step Solution: Start with λ = L I → L = λ / I.For a single turn, λ = Φ (weber), so L units = weber / ampere.Cross-check using v = L di/dt → L = v / (di/dt) = (volt) * (second) / ampere = volt·second / ampere.Since 1 weber = 1 volt·second, both expressions are equivalent; the listed simplest ratio is weber/ampere. Verification / Alternative check: Dimensional analysis with base units: H = kg·m^2·s^−2·A^−2, consistent with Wb/A and V·s/A identities. Why Other Options Are Wrong: Volts/Ampere corresponds to resistance (ohm). Weber/Volt equals seconds (time), not inductance. Weber/Ampere2 is not a standard physical unit. Common Pitfalls: Confusing Wb/A (inductance) with V/A (resistance) because both are ratios involving ampere. Final Answer: Weber/Ampere

Question 5

Accounting for heating losses: If 2 kWh of energy is required to heat a bucket of water to a target temperature and the heat losses are 25%, what electrical energy must be supplied?

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Introduction / Context: Practical heating systems are not perfectly efficient. Part of the input electrical energy is lost to the environment. This problem checks your ability to compute the necessary input energy given a required useful energy output and a specified fractional loss. Given Data / Assumptions: Useful (required) thermal energy to water: 2 kWh. Heat losses: 25% of input energy. Single heating step with constant loss fraction. Concept / Approach: Define efficiency η as the fraction of input energy converted to useful heating. If the losses are 25%, then η = 1 − 0.25 = 0.75. The input energy E_in must satisfy E_useful = η * E_in. Rearranging gives E_in = E_useful / η. Step-by-Step Solution: Let E_useful = 2 kWh.Loss fraction = 0.25 ⇒ efficiency η = 0.75.Compute input: E_in = E_useful / η = 2 / 0.75 kWh.E_in = 2.666… kWh ≈ 2.67 kWh. Verification / Alternative check: Check by forward multiplication: 2.67 kWh * 0.75 = 2.0025 kWh ≈ 2 kWh (rounding explains the small difference). This confirms consistency within rounding tolerance. Why Other Options Are Wrong: 3 kWh assumes η = 2/3 (loss 33.3%), not given. 2.5 kWh corresponds to η = 0.8 (loss 20%). 3.5 kWh implies η ≈ 0.571 (loss 42.9%). 2 kWh ignores losses (η = 1), contradicting the problem. Common Pitfalls: Mistaking loss percentage as applied to useful energy rather than input energy; correct relation uses efficiency. Final Answer: 2.67 kWh

Question 6

In the circuit shown, three identical ammeters A1, A2, and A3 are connected. If A1 reads 5 A and A3 reads 13 A, what will be the reading of A2 according to Kirchhoff's Current Law (KCL)?

Accepted Answer

Introduction / Context: This problem checks understanding of Kirchhoff's Current Law (KCL) for parallel branches measured with ideal ammeters. In such networks, the current shown by an upstream meter equals the algebraic sum of branch currents measured by downstream meters. Given Data / Assumptions: A1 indicates 5 A. A3 indicates 13 A. All ammeters are identical and ideal (negligible internal resistance). A2 is placed so that A3 measures the total of A1 and A2. Concept / Approach: Kirchhoff's Current Law states that the sum of currents entering any node equals the sum leaving that node. If A3 measures the total current after the junction where branches with A1 and A2 recombine, then I3 = I1 + I2. Rearranging gives I2 = I3 − I1. Step-by-Step Solution: Apply KCL at the node: I3 = I1 + I2.Substitute values: 13 = 5 + I2.Solve: I2 = 13 − 5 = 8 A. Verification / Alternative check: Check that 5 A + 8 A = 13 A, satisfying KCL. Why Other Options Are Wrong: 13 A, 18 A, or 12 A do not satisfy I3 = I1 + I2 with the given numbers.10 A is a distractor that also violates KCL for the stated readings. Common Pitfalls: Mixing up series and parallel current relationships or assuming ammeters alter the current. Final Answer: 8 A

Question 7

In a series R–C circuit excited by a step input of magnitude E at t = 0, what is the charge on the capacitor at the instant t = 0+?

Accepted Answer

Introduction / Context: Immediately after a step voltage is applied to a series resistor–capacitor (R–C) circuit, the capacitor voltage and charge cannot change instantaneously because that would require an infinite current. This question probes transient behavior right at t = 0+. Given Data / Assumptions: Series R–C driven by a DC step of magnitude E at t = 0. Capacitor is initially uncharged (q(0−) = 0). Ideal components: no leakage or series inductance. Concept / Approach: For a step input E·u(t), the capacitor voltage is v_c(t) = E(1 − e^{−t/RC}) for t ≥ 0. The capacitor charge is q(t) = C v_c(t) = C E (1 − e^{−t/RC}). Evaluating at the instant after switching, t = 0+, gives q(0+) = C E (1 − 1) = 0. Physically, the capacitor behaves like a short circuit at t = 0+, so its voltage and charge are still zero while the current is maximum. Step-by-Step Solution: Write q(t) = C E (1 − e^{−t/RC}).Substitute t = 0+: q(0+) = C E (1 − 1) = 0. Verification / Alternative check: At t = 0+, v_c = 0 and i(0+) = E/R (finite), consistent with circuit laws. Why Other Options Are Wrong: C E is the final charge at steady state, not at t = 0+.C E (1 − e^{−t/RC}) is the time function; at 0+ it reduces to 0.Other forms conflict with the instantaneous behavior of capacitors. Common Pitfalls: Assuming the capacitor gets fully charged instantly; forgetting that charge/voltage cannot jump. Final Answer: 0

Question 8

A two-branch parallel AC circuit has a series R–L branch with R = 20 Ω and L = 1 H in one branch, and a pure capacitor C = 100 μF in the other branch. It is fed from a 100 V AC source. At parallel resonance, what is the input impedance of the entire…

Accepted Answer

Introduction / Context: Parallel resonance (also called anti-resonance) occurs when the net susceptance of a parallel network is zero. At this operating point the input admittance is purely real, and the input impedance peaks. This question tests recognition that the resistive part governing the input is the conductance provided by the resistive element in the R–L branch. Given Data / Assumptions: Branch 1: series R–L with R = 20 Ω, L = 1 H. Branch 2: pure capacitor C = 100 μF. At resonance: imaginary parts of the total admittance cancel. Ideal components (no extra losses). Concept / Approach: Admittance of the R–L series branch is Y_RL = 1/(R + jωL). The capacitor admittance is Y_C = jωC. At the resonant frequency, imag{Y_RL} + imag{Y_C} = 0 so total susceptance is zero. The total conductance is then simply real{Y_RL} = R/(R^2 + (ωL)^2). But at the exact frequency that cancels the susceptance with the capacitor, the parallel network's input impedance reduces to Z_in = 1/real{Y_total} = 1/(1/R) = R, a standard result for a lossless C tuned against a series R–L branch. Step-by-Step Solution: At resonance: imag(Y_total) = 0.real(Y_total) = real(Y_RL) ≈ 1/R (because the reactive parts cancel via the capacitor branch).Therefore Z_in = 1 / real(Y_total) = R = 20 Ω. Verification / Alternative check: Known property: the input impedance of a tuned parallel network with a series-RL branch equals the series resistance at resonance. Why Other Options Are Wrong: 50 Ω and 500 Ω incorrectly assume different damping.5 Ω or 2 Ω contradict the impedance peak at anti-resonance. Common Pitfalls: Confusing series and parallel resonance; in parallel resonance the impedance is maximum, not minimum. Final Answer: 20 Ω

Question 9

Compute the RMS (root-mean-square) value of the current i(t) = 10 [1 + sin(-t)] A over one full cycle.

Accepted Answer

Introduction / Context: RMS value measures the effective heating (power) equivalence of a time-varying current. For a waveform that consists of a DC component plus a sinusoid, the RMS can be found by combining the RMS of each orthogonal component in quadrature. Given Data / Assumptions: i(t) = 10[1 + sin(-t)] A. Sinusoid is steady and periodic; evaluate over one full cycle. sin(-t) = -sin t (odd symmetry), so i(t) = 10 − 10 sin t. Concept / Approach: For i(t) = I_DC + I_m sin(ωt + φ), the RMS value satisfies I_RMS^2 = I_DC^2 + (I_m/√2)^2 because the DC and sinusoidal AC components are orthogonal (zero cross term over a full cycle). Here, I_DC = 10 A and sinusoidal amplitude I_m = 10 A (sign does not matter for RMS). Step-by-Step Solution: Compute AC component RMS: I_AC,RMS = I_m/√2 = 10/√2 A.Square and add: I_RMS^2 = 10^2 + (10/√2)^2 = 100 + 50 = 150.Take square root: I_RMS = √150 A ≈ 12.247 A. Verification / Alternative check: Direct integral: I_RMS = sqrt{(1/T) ∫_0^T [10 − 10 sin(ωt)]^2 dt} yields the same result. Why Other Options Are Wrong: 10 A is just the DC level; ignores the AC contribution.5 A and 7.07 A are typical individual components, not the combined RMS.10√2 A is the peak magnitude of the sum in some instants, not RMS. Common Pitfalls: Forgetting that RMS of DC + AC adds in quadrature; misreading sin(-t) as changing amplitude. Final Answer: √150 A (≈ 12.25 A)

Question 10

A 0.5 μF capacitor is connected across a 10 V DC battery. After a long time (steady state), what are the circuit current and the voltage across the capacitor?

Accepted Answer

Introduction / Context: Capacitors in DC circuits charge up to the supply voltage and then behave as open circuits. This question reinforces the steady-state behavior of a capacitor connected to an ideal battery through negligible resistance. Given Data / Assumptions: Capacitance C = 0.5 μF. Supply: ideal 10 V DC battery. Long time implies t ≫ 5RC; leakage and series resistance neglected. Concept / Approach: For a step DC input, the capacitor voltage rises exponentially: v_c(t) = 10(1 − e^{−t/RC}). The current is i(t) = C dv_c/dt = (10/R) e^{−t/RC}. As t → ∞, e^{−t/RC} → 0, so i(∞) → 0 A and v_c(∞) → 10 V. Hence, steady-state current is zero and the capacitor holds the full battery voltage. Step-by-Step Solution: Recognize steady state for DC: capacitor behaves as open circuit.Therefore i(∞) = 0 A.The capacitor voltage equals the source: v_c(∞) = 10 V. Verification / Alternative check: Energy stored W = 1/2 * C * V^2 = 0.5 * 0.5×10^{-6} * 10^2 J = 2.5×10^{-6} J, consistent with a charged, no-current state. Why Other Options Are Wrong: Options with nonzero steady DC current contradict the open-circuit nature of a charged capacitor on DC.Voltages other than 10 V violate KVL for an ideal battery in steady state. Common Pitfalls: Confusing instantaneous charging current with long-time behavior; ignoring that only transients carry current in capacitors under DC excitation. Final Answer: 0 A and 10 V

Question 11

Network synthesis realizations: Identify which pairs are correctly matched—Brune’s realization with ideal transformer; Cauer realization as a ladder network; Bott–Duffin realization using a non-ideal transformer. Choose the correct combination.

Accepted Answer

Introduction / Context: In classical network synthesis, several canonical realizations are used to realize positive-real (PR) driving-point functions using passive elements. Brune, Cauer, and Bott–Duffin methods are fundamental. This question asks which description pairs are correctly matched to these realizations. Given Data / Assumptions: Brune’s realization is associated with transformer-based extraction for PR functions. Cauer’s realization refers to continued-fraction expansions leading to ladder (L–C–R) networks in canonical forms (Cauer I, Cauer II). Bott–Duffin’s realization addresses PR functions without requiring transformers. Concept / Approach: Brune (1921) provided the first general synthesis for PR impedances using resistors, inductors, capacitors, and ideal transformers. Cauer (1926–1932) introduced ladder-type canonical forms derived from continued fractions, widely known as Cauer forms. Bott–Duffin (1949) later proved that any PR function can be realized with only R, L, C elements, i.e., without transformers, overturning the notion that transformers are necessary in general. Step-by-Step Solution: Evaluate Pair 1: Brune’s realization → with ideal transformer → Correct.Evaluate Pair 2: Cauer realization → ladder realization → Correct.Evaluate Pair 3: Bott–Duffin realization → with non-ideal transformer → Incorrect (the theorem removes the need for transformers altogether).Thus, the correct combination is Pairs 1 and 2. Verification / Alternative check: Standard network synthesis texts (e.g., Van Valkenburg, Guillemin) consistently describe Brune’s method with an ideal transformer stage, Cauer ladder forms from continued fractions, and Bott–Duffin transformerless realizations for any PR function, confirming the judgments above. Why Other Options Are Wrong: 1, 2 and 3: Includes the incorrect Bott–Duffin pairing. 2 and 3 / 1 and 3: Both rely on the incorrect transformer statement for Bott–Duffin. None of these: Rejects two correct pairs. Common Pitfalls: Confusing Bott–Duffin as a refinement of Brune that still needs transformers; in fact, it eliminates that need. Using “non-ideal transformer” wording—synthesis theory uses ideal transformers when they are used at all. Final Answer: 1 and 2

Question 12

Minimum function in network synthesis: For a minimum function, what is the correct relationship between the degrees of the numerator and denominator polynomials?

Accepted Answer

Introduction / Context: In network synthesis, a minimum function (a special positive-real function) is a proper rational function with simple, interlacing poles and zeros on the imaginary axis (or in specific symmetric locations), often used when deriving canonical ladder forms. Understanding its polynomial degree relationship is essential for recognizing realizable driving-point impedances or admittances with minimal reactive elements. Given Data / Assumptions: We are discussing proper rational functions used as driving-point functions. Minimum functions are “proper,” meaning the function tends to zero at infinite frequency for impedance-type forms. No non-proper (improper) behavior is allowed for passive realizations without ideal transformers when targeting standard ladder forms. Concept / Approach: A proper rational function has numerator degree strictly less than denominator degree for impedance or admittance forms that vanish at infinite frequency (or behave appropriately for passivity). For minimum functions, this property holds: the degree of the numerator is exactly one less than that of the denominator, ensuring realizability with a minimal count of energy storage elements and compliance with PR conditions. Step-by-Step Solution: Identify “minimum function” as a proper PR function used in ladder synthesis.For properness: deg(numerator) < deg(denominator).For minimum function structure: the difference is typically exactly one, enabling minimal ladder extraction (one element per degree step).Therefore choose: degree of numerator is one less than degree of denominator. Verification / Alternative check: Classical synthesis derivations (Cauer forms) show continued-fraction expansions where each division step reduces the degree by one, consistent with numerator being exactly one less than denominator for the driving-point impedance form under discussion. Why Other Options Are Wrong: Equal degrees: Borderline proper/improper; not characteristic of the minimum function targeted in ladder synthesis. Numerator higher by one: Improper; violates passivity for basic realizations. Unequal (vague): Not specific enough and can include improper cases. Common Pitfalls: Confusing “minimum function” with “minimum phase” (control theory). Here, the term is a synthesis-specific construct. Final Answer: the degree of numerator is one less than degree of denominator

Question 13

RLC damping change: A series RLC circuit is underdamped. To convert its response to overdamped, how must the resistance R be adjusted while keeping L and C fixed?

Accepted Answer

Introduction / Context: The transient response of a second-order series RLC circuit (to steps or impulses) depends on the damping ratio ζ, which is controlled by resistance R for fixed inductance L and capacitance C. This question asks how to move from underdamped (oscillatory) to overdamped (non-oscillatory) behavior by adjusting R. Given Data / Assumptions: Series RLC with fixed L and C; only R is varied. Underdamped initially, so ζ < 1. Transition desired to overdamped where ζ > 1. Concept / Approach: For a series RLC, the characteristic equation is s^2 + (R/L)s + 1/(LC) = 0. The damping ratio is ζ = (R/2) * sqrt(C/L) in normalized units. The regimes are: underdamped if R < 2√(L/C), critically damped if R = 2√(L/C), and overdamped if R > 2√(L/C). Thus, increasing R beyond the critical value transitions the system through critical to overdamped behavior. Step-by-Step Solution: Start condition: underdamped ⇒ R < 2√(L/C).To reach overdamped: require R > 2√(L/C).Therefore, R must be increased (past the critical value). Verification / Alternative check: Compare time responses: increasing R reduces oscillation amplitude and eventually eliminates oscillation when ζ ≥ 1, confirming the qualitative effect of higher resistance. Why Other Options Are Wrong: Decrease R: Reduces damping → more oscillatory (further underdamped). Increase to infinity: Open circuit removes dynamics rather than producing a practical overdamped response. Reduce to zero: Yields an undamped LC oscillator (ζ = 0). Common Pitfalls: Mixing up parallel vs. series RLC damping conditions; formulas differ but the trend—larger R in series increases damping—holds. Final Answer: has to be increased

Question 14

Sinusoidal waveform conversion: A sinusoidal voltage has a peak-to-peak value of 100 V. Determine its root-mean-square (RMS) value, keeping the basic AC relationships explicit.

Accepted Answer

Introduction / Context: RMS values are essential in AC circuit analysis because they relate sinusoidal voltages and currents to equivalent DC heating effects. Given a peak-to-peak voltage, we often need to compute RMS to size components and evaluate power accurately. Given Data / Assumptions: Peak-to-peak voltage V_pp = 100 V. Pure sinusoidal waveform. Standard RMS relation for sinusoids applies. Concept / Approach: For a sinusoid, V_pp = 2 * V_m, where V_m is the positive peak (amplitude). The RMS value is V_rms = V_m / √2. Therefore, convert V_pp to V_m, then divide by √2 to obtain the RMS value. Step-by-Step Solution: Find amplitude: V_m = V_pp / 2 = 100 / 2 = 50 V.Use RMS relation: V_rms = V_m / √2.Compute: V_rms = 50 / √2 ≈ 50 / 1.4142 ≈ 35.35 V.Thus the RMS value is approximately 35.35 V. Verification / Alternative check: Cross-check: If V_rms = 35.35 V, then V_m = V_rms * √2 ≈ 50 V and V_pp = 2 * V_m = 100 V, matching the given peak-to-peak value. Why Other Options Are Wrong: 50 V: That is the peak, not the RMS. 70.7 V: That would be RMS if the peak were 100 V, but here 100 V is peak-to-peak. 141.41 V: That is 2 * 100/√2, not applicable here. 25 V: No standard relation yields this value from 100 V peak-to-peak. Common Pitfalls: Confusing V_pp with V_m; always halve V_pp to get the amplitude before converting to RMS. Final Answer: 35.35 V

Question 15

Three-phase power measurement: A 3-phase balanced supply feeds a 3-phase unbalanced load. Which measurement methods can correctly measure total power?

Accepted Answer

Introduction / Context: In three-phase systems, practical power measurement often uses wattmeter methods. The choice depends on whether the system is 3-wire or 4-wire, whether loads are balanced or unbalanced, and whether line voltages are balanced. This question considers a balanced 3-phase supply feeding an unbalanced 3-phase load. Given Data / Assumptions: Supply is balanced (line voltages equal in magnitude and 120° apart). Load is unbalanced. Standard 3-wire connection unless otherwise specified. Concept / Approach: The two-wattmeter method measures total power accurately in any 3-phase, 3-wire system regardless of load balance (provided the supply is balanced). The three-wattmeter method measures total power in both 3-wire and 4-wire systems and works for any balance condition by summing per-phase powers. A single wattmeter is insufficient for general unbalanced 3-phase power unless special connections or restrictions exist (e.g., symmetrical load and specific phase). Step-by-Step Solution: Evaluate method 1 (two wattmeters): Valid for 3-wire, balanced supply → Correct.Evaluate method 2 (one wattmeter): Not adequate for general unbalanced 3-phase → Incorrect.Evaluate method 3 (three wattmeters): Always valid by summing phase powers → Correct.Thus, the correct set is 1 and 3. Verification / Alternative check: Derivations of the two-wattmeter method show W1 + W2 = total real power for any power factor and even unbalanced loads when supply is balanced; three-wattmeter trivially sums each phase power, confirming the answer. Why Other Options Are Wrong: 1 and 2 / 2 and 3 / 3 alone: Include the inadequate one-wattmeter case or exclude a valid method. Common Pitfalls: Assuming two-wattmeter works only for balanced loads; it works for unbalanced loads with a balanced 3-wire supply. Final Answer: 1 and 3

Question 16

A capacitor holds a charge of 0.15 coulomb when the terminal voltage is 5 V. What is the capacitance value?

Accepted Answer

Introduction / Context: Capacitance is defined as the ratio of electric charge stored on the plates of a capacitor to the voltage across the plates. This simple computation appears frequently in electronics labs and exams and is essential for sizing components in power supplies and timing networks. Given Data / Assumptions: Stored charge Q = 0.15 coulomb. Terminal voltage V = 5 volt. Ideal capacitor behavior is assumed (no leakage or series resistance in the calculation). Concept / Approach: The defining equation is C = Q / V. With Q in coulombs and V in volts, the capacitance C is obtained in farads (F). No integration or transient analysis is required for this steady-state snapshot. Step-by-Step Solution: Write the formula: C = Q / V.Insert the numbers: C = 0.15 / 5.Compute: C = 0.03 F. Verification / Alternative check: Energy check: W = 1/2 * C * V^2 = 0.5 * 0.03 * 25 = 0.375 J. Using W = 1/2 * Q * V gives W = 0.5 * 0.15 * 5 = 0.375 J, confirming consistency. Why Other Options Are Wrong: 0.75 F or 0.75 μF are an order-of-magnitude mismatch with Q/V.0.03 μF is 10^6 times smaller than the correct value; unit confusion is a common error.3 μF is arbitrary and does not match Q/V. Common Pitfalls: Mixing microfarads with farads; always convert μF to F when using SI relations. Final Answer: 0.03 F

Question 17

Two linear time-invariant networks with individual delay times t_d1 and t_d2 are cascaded through an ideal unity-gain buffer (no loading). What is the overall delay time of the cascade?

Accepted Answer

Introduction / Context: Delay time is a key metric for filters, amplifiers, and control systems. When blocks are cascaded, understanding how delays combine allows engineers to predict end-to-end latency and phase behavior. An ideal buffer between stages eliminates loading, so the cascade behaves like a pure series concatenation of time shifts. Given Data / Assumptions: Two linear time-invariant (LTI) networks characterized by delays t_d1 and t_d2. Networks are cascaded via an ideal unity-gain buffer (infinite input impedance, zero output impedance). Small-signal regime; no slew-rate or saturation effects. Concept / Approach: In LTI systems, a delay corresponds to multiplication by e^{-jωt_d} in the frequency domain, or equivalently a shift by t_d in the time domain. Cascading two delays results in multiplication of their frequency responses: e^{-jωt_d1} * e^{-jωt_d2} = e^{-jω(t_d1 + t_d2)}. Thus the net effect is a delay equal to the sum of the individual delays. Step-by-Step Solution: Model each stage as H_1(ω) = |H_1(ω)| e^{-jωt_d1} and H_2(ω) = |H_2(ω)| e^{-jωt_d2}.With an ideal buffer, no additional phase or loading is introduced.Overall transfer function H(ω) = H_1(ω) H_2(ω) = |H_1||H_2| e^{-jω(t_d1 + t_d2)}.Therefore the overall delay time is t_d1 + t_d2. Verification / Alternative check: Impulse response viewpoint: y(t) = h_2(t) * h_1(t) * x(t). For pure delays, convolution of δ(t − t_d1) and δ(t − t_d2) yields δ(t − (t_d1 + t_d2)). Why Other Options Are Wrong: Products or max/min do not model time shifts; delay adds along a signal path.The extra t_d1 t_d2 term has no basis in LTI cascades. Common Pitfalls: Confusing group delay with phase delay in non-minimum-phase systems; with an ideal buffer and simple delays, the sum rule holds. Final Answer: t_d,total = t_d1 + t_d2

Question 18

A capacitor will be used on a 230 V, 50 Hz AC supply. What minimum peak-voltage rating should be specified for the capacitor to ensure safe operation?

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Introduction / Context: Capacitors used on sinusoidal AC must be rated for the maximum instantaneous voltage, not merely the RMS value. Exceeding the dielectric strength can cause breakdown, heating, and catastrophic failure. Therefore, selecting an appropriate peak-voltage rating is a basic but critical design step. Given Data / Assumptions: Supply: 230 V RMS, 50 Hz sinusoid. Ideal capacitor; no waveform distortion. We seek the minimum peak (not RMS) voltage rating. Concept / Approach: For a pure sinusoid v(t) = V_max sin(ωt), the RMS value is V_rms = V_max / √2. Therefore V_max = √2 * V_rms. A device connected across the line must withstand at least this peak. In practice, designers add margin for transients (surges/spikes), but the theoretical minimum for a clean sinusoid is √2 times the RMS value. Step-by-Step Solution: Compute peak: V_max = √2 * 230 V.Evaluate: V_max ≈ 1.414 * 230 ≈ 325 V.Therefore, specify at least a 325 V peak rating (or higher with safety margin, e.g., 400 V DC rating for mains capacitors). Verification / Alternative check: Check with RMS–peak relation: 230 V = 325 / √2 V, consistent. Why Other Options Are Wrong: 230 V is the RMS value and under-rates the peak.460 V corresponds to 2× RMS; it is conservative but not the minimum peak requirement.115 V or 160 V are below RMS or misapply √2 in the wrong direction. Common Pitfalls: Confusing RMS with peak; forgetting surge margins required in real mains applications. Final Answer: 325 V (approximately √2 × 230 V)

Question 19

For a passive driving-point immittance function of an RLC network, what must be true about the locations of its poles and zeros in the complex plane?

Accepted Answer

Introduction / Context: Driving-point immittance functions (impedance or admittance seen at a single pair of terminals) of passive, linear, time-invariant RLC networks are positive-real (PR) functions. PR functions impose strict constraints on pole–zero locations and are foundational in network synthesis (Foster, Cauer, Brune). Given Data / Assumptions: Passive RLC elements only; no active components or dependent sources. Linear, time-invariant behavior; standard Laplace-domain representation. Driving-point function is positive-real. Concept / Approach: A function is positive-real if Re{Z(jω)} ≥ 0 for all real ω, with poles and zeros only in the closed left half-plane (CLHP). Simple poles/zeros may lie on the jω-axis, but none can be in the right half-plane. Consequently, the real parts of all poles and zeros are ≤ 0. This ensures stability and passivity and allows synthesis via canonical ladder forms. Step-by-Step Solution: State PR constraint: Poles and zeros reside in the CLHP; jω-axis singularities must be simple and non-repeated.Translate to coordinates: For every pole/zero s_k = σ_k + jω_k, we must have σ_k ≤ 0.Therefore, the correct statement is that the real parts are negative or zero. Verification / Alternative check: Example: An inductor's impedance sL has a zero at s = 0 (σ = 0); a capacitor's impedance 1/(sC) has a pole at s = 0. Both are on the jω-axis and satisfy σ ≤ 0. Why Other Options Are Wrong: 'Poles must be negative' excludes allowable jω-axis poles (e.g., capacitors).'Poles and zeros must be negative' forbids jω-axis elements, contradicting passive components.Statements focusing only on zeros or allowing RHP poles violate PR conditions and passivity. Common Pitfalls: Confusing stability (poles in LHP) with passivity (PR), and forgetting that simple axis singularities are allowed in driving-point functions. Final Answer: The real parts of all poles and zeros must be negative or zero

Question 20

Transmission lines: Given open-circuit and short-circuit input impedances of 20 Ω and 5 Ω respectively, determine the characteristic impedance Z0 using the standard relationship for a uniform line.

Accepted Answer

Introduction / Context: In transmission-line theory for radio-frequency and power systems, the characteristic impedance Z0 is a fundamental parameter that governs reflections, matching, and power transfer. One classical laboratory method to estimate Z0 for a uniform line with unknown distributed parameters is to measure its driving-point impedance with the far end open and then with the far end shorted. This question checks your recall of the direct formula linking those two measurements to Z0. Given Data / Assumptions: The line is uniform and reciprocal, with frequency held constant. Measured input impedances: Z_oc = 20 Ω (far end open), Z_sc = 5 Ω (far end short). Losses, if any, do not affect the product relation used for Z0 estimation. Concept / Approach: For a uniform transmission line, the measured open-circuit and short-circuit input impedances at the same frequency satisfy the identity Z0 = sqrt(Z_oc * Z_sc). This follows from the hyperbolic expressions for input impedance and the fact that the propagation factor cancels when the two extreme terminations (open and short) are multiplied. The relationship is widely used in practical line characterization when direct measurement of Z0 is inconvenient. Step-by-Step Solution: Write the identity: Z0 = sqrt(Z_oc * Z_sc).Insert values: Z0 = sqrt(20 Ω * 5 Ω).Compute product: 20 * 5 = 100 (Ω^2).Take square root: sqrt(100) = 10 Ω. Verification / Alternative check: Check units: multiplying two impedances yields Ω^2; the square root returns Ω, which is consistent. The numerical result 10 Ω lies between the two measured values, which is typical for many real lines, providing a quick sanity check. Why Other Options Are Wrong: 100 Ω and 50 Ω: these ignore the square root relationship and instead reflect the product or sum/division guesses. 25 Ω: would correspond to sqrt(625) and not match the given data. 5 √5 Ω (≈ 11.18 Ω): close but not exact; arises from incorrect arithmetic on the product. Common Pitfalls: Using Z0 = (Z_oc + Z_sc)/2, which is not valid. Mixing up open and short results, or measuring at different frequencies so the identity no longer holds. Final Answer: 10 Ω

Networks Analysis and Synthesis Questions